100 WATT UL AB1 MONO BLOC  AMPLIFIERS
Content of this page :-
Picture of  two 100W UL-AB1 mono amps, general description,
Schematics for :- 1 power amp schematic, 2 power supply schematic, 3 bias and protection circuits.
graphs for :- 4 power vs load graphs, 5 THD graphs. Full explanations of all included.



I sold these two amplifiers in early 2004 to a delighted customer living near Cairns with a tropical climate.
He wished to use them to obtain suitably enthralling performance with Quad ESL63 speakers used mainly during the dry season. The music with these amps has rock solid bass, the usual clear mids, and well detailed treble, with an overall ease and precision of a high amount of class A.
During the wet season such amps run well, with the warmth from the tubes tending to keep the amps free of moisture.
They have aluminium chassis, 470 mm long x 260 mm wide x 220 high, and a protective grille over the tubes of steel rod.
There is no cover needed over the other chassis mounted items. Weight is approx 22 Kg, ( 49 Lb ), for each mono amp.
The largest transformer is the output transformer with a 75 mm stack of 44 mm tongue E&I silicon steel lams, seen just behind the output tubes. The power transformer is 100 mm stack of 38 mm iron, seen furthest from view.
Silicon rectifiers are used, with C-L-C filtering using a choke of 2H and large value electrolytics, to maintain an impeccably low noise level.
The amps in the above image have 6 x Sovtek EL34WXT. There was provision for a pair of paralled 12AX7 input tubes which were able to be used instead of a single 12AU7 or 12AT7 which may otherwise be used
and without having to change any circuitry to take the different tubes.
But after many trials, I decided that there were no sonic benefits to be had by changing input tubes and I finally settled on one 12AU7. The amp was sold in 2004 with matched 6CA7EH which I feel are more rugged than EL34, and excellent sounding.

Future 100W mono blocs  will be using  a quad of 6550, KT88, KT90 instead of a six pack of EL34/6CA7, but will have an almost identical excellent performance, and in 2004 I had a large batch of steel chassis made up for the next models.
 
 
SCHEMATICS AND TECHNICAL NOTES :-
Sheet 1, 100w UL AB1 power amp schematic.
Sheet 2, 100w UL AB1 power supply schematic.
Sheet 3, 100w UL AB1 bias and protection circuits.
Sheet 4, 100w UL AB1  power vs load graphs.
Sheet 5, 100w UL AB1 THD graphs.
These schematics are of the 100w mono amps as supplied to my customer in 2004, and supersede the first version drawn up for the 2000 edition of the website.
 
 
Sheet1 100w UL AB1 power amp.

 
Fixed bias is used, and there are six adjust pots and volt meter test points accessible through 6 access holes in the side of the chassis, so bias can be adjusted easily with a cheap dc voltmeter and screw driver without moving the amps to a work bench or removing any covers.
Driver tubes are 2 x NOS 6SN7 with both triodes in each paralleled. This pair of tubes are set up in a differential amp or long tail pair driver/phase inverter with  their commoned cathodes connected to a "long tail" using one MJE340, which has such a
high collector dynamic impedance in the circuit that it can be regarded as a constant current source.
This makes the two output opposite phased outputs from the 6SN7 tubes exactly equal in amplitude which depends solely
on the equality of their resistor loads. The MJE340, has no sonic signature, and does no active amplification and acts merely as a high ac impedance source of cathode current and it replaces what would be pentode tube which would work no better.
Input tube is 12AU7 with both triodes paralleled. The output of the input triode feeds one of the grids of the LTP via a LF gain shelving network to tailor the open loop gain and phase response to ensure unconditional stablity when global
NFB is applied. The 47nF + 1M in series with the LTP grid cause an open loop loss of gain below 20 Hz, but extend the final pole of LF determined by the 0.47 uF and 1M + 220k grid bias resistor.
 The effect of this measure is superb LF stability.
Series R + C networks are also used across each half primary, 1.5k + 0.0022uF, and the secondary of the output transformer, 0.22uF + 15 ohms to ensure the amp is loaded by a resistive load at HF, when typically inductive speaker loads are used.
The traditional global negative voltage FB resistor of 2 x 2.7k have 1,000pF across them to advance the phase of the global NFB voltage signal applied from the positive speaker terminal to the V1 cathode circuit. The 1,000pF compensates for the phase lag in the open loop phase response.
There is also some global small amount of negative current feedback applied from the negative speaker terminal and taken to the V1 cathode circuit point above the 2uH and 0.1 ohm LR network.
Above 20 kHz, NCFB applies itself to prevent ringing in square wave responses
with capacitor loads. These measures ensure complete freedom from any oscillation with any load or no load at all.
The seriesed silicon diodes from the output anode to 0V are to limit the peak output voltage swing to the 430 volt supply voltage, since these diodes conduct when either side of the primary tries to swing to a negative value, caused by back emf phenomena in the OPT when no load is present, and a high output voltage is used. This measure safeguards stressing the OPT insulation by excessive voltages.
Power output can be seen plotted in the the sheet 4, for selectable triode or UL connected amps.
Bandwidth at 72 watts, 8 ohms, is 10 Hz to 65 kHz, output impedance < 0.5 ohms,
Distortion < 0.3% at 100 watts, 4 ohms, and less than 0.05% at 3 watts.
Noise is very low.
The amp is fitted with the usual active protection circuits, shown on sheet 3.
An led on the front of the chassis indicates clipping, or any fault condition.
 
Sheet 2, 100W UL AB1 amp power supply


The power supply has all solid state rectifiers. A CLC filter ensures low B+ rail noise and a 50 ohm resistor limits peak charge currents into the two 220uF series input caps to the CLC for B+.
The 50 ohms also keeps the B+ at about +410V to enable a high idle bias current
thus ensuring a large amount of class A power. DC is applied to V1 heaters and fixed grid bias and B+ supplied to V1 is shunt
regulated. There is a relay in the power transformer HT secondary circuit which is operated by the protection circuit
shown in sheet 3.
 
Sheet 3,  100W UL AB1 bias and protection schematic.


The fixed bias adjustment pots are arranged as shown to give each tube a range of applied grid bias between -33V and -48V, and each pot for each output tube grid bias is adjusted so 0.5Vdc appears across the same tube's 10 ohm cathode resistor. This indicates 50mA of idle current.
The process of setting the bias for the amp is repeated 3 times after the amp has warmed up since bias adjustments are interactive.
The circuit is actively protected against bias failure in one or more tubes, or against excessive and continued use with loads which are too low in value.
Each cathode of each output tube is grounded through 10 ohms and points K1 to K6 are all fed to a common cathode monitoring signal path which has its voltage reduced by the R divider of 4k7 and 2k7 with a 220uF cap to remove unwanted ac signals during operation.
Should one of more cathode dc currents rise to dangerous levels for more than a couple of seconds, the scr will trip and  the 16V relay supply will be pulled to 0V since the 50 ohm 10W resistor is grounded at the relay end.
There is a 6 second delay for turn on of the B+.
If the scr is tripped the fault led will turn on. The same led will flash if the amp clips since the error signal from the output of V1 is fed through a high resistance path to a darlington pair of bjt driving a second gain bjt to turn on an led.
All the protection parts will fit on a small board under the chassis and all can be easily sourced at any electronic parts shops.
 
Sheet 4, power vs load graph.



 The graphs show maximum power levels at thd <2% just before clipping with a range of load values.
Curves A and B are for Ultralinear with either 2k : 6 ohm load match or 4.5k : 6 ohms.
Curves C and D are for Triode with either 2K : 6 ohms or 4k5 : 6 ohms.
Any value of load along the RL bottom line can be chosen and the output power max can be seen on the curves.

Using the higher Z ratio on the OPT results in less maximum  output power for most common loads between 4 and 8 ohms but the class A portion of total power increases and although the total AB power reduces thethd and output impedance
are both much lower.

For example, suppose we have the OPT set for a load match for 4k5 : 6 and we connect 8 ohms,
then the maximum UL power output only 52 watts, but it is virtually all class A and the thd will be about 1/2 that of 6 ohms.

Load matching is all very confusing to most people.
A common misunderstanding is that the more ohms a speaker has, the more difficult it is to drive.
But more ohms means a less difficult load. Carrying 10 bricks is more difficult than carrying 1 brick.
But like so many things in electronics basic commonsense seems reversed to a novice. If anyone cannot understand
they need may like to study more about the basics of tube operation and load matching effects and distortions
in such wonderful books such as the 1955 4th Ed of Radiotron Designer's Handbook, or read my pages on basic tube use issues.

Sheet 5, 100W UL AB1 THD figures for two amps.

 
The graphs of  total harmonic distortion, thd, are for one pair of 100 watt tube amps with identical schematics but despite this there are two curves for amps A and B and there is up to 7dB difference between thd at below 1 watt. The graphs were made during completion work on the two amps. The voltage scale is linear and the THD scale is *logarithmic* to display small quantities of THD more easily. The amp A had the highest THD which is listed below with power levels and what
would be SPLs using  average modern speakers of 6 ohms and 90dB/W/M :-

1 watt , 2.45vrms, 0.02%, 90dB SPL,
2 watts, 3.46Vrms, 0.03%, 93dB SPL,
4 watts, 4.89Vrms, 0.047%, 96dB SPL,
8 watts, 6.93Vrms, 0.09%, 99dB SPL,
16 watts, 9.8Vrms, 0.15%, 102dB SPL,
32 watts, 13.8Vrms, 0.20%, 105dB SPL,
64 watts, 19.6Vrms, 0.37%, 108dB SPL,
88 watts, 23.0Vrms, 1.0% 109dB SPL, and DEAFENINGLY LOUD!

Like nearly all amps the fidelity increases with a higher load value but with a reduction of maximum output power.
Most people will find that with two amplifiers average levels of 1/2 a watt from each will produce SPLs of 88 dB approximately although peaks in drumbeats and transients will go a lot higher but these will be very easily dealt with by these amps. The average level for most loud music in a domestic situation is 88db for men, and 84dB for women, and that includes both channels, so indeed these amps have quite enough power even for teenagers who like bass boosting if possible in the preamp.
( OK, you have a killer teenager?... I don't want to know..)

Much is said about tube amp distortions spoiling performances but let's get this into perspective.
If the speaker voltage is 5Vrms at 4 watts of level, then the THD = 0.05%. Therefore the actual distortion voltage within the signal is 0.0025Vrms, and would be very difficult to hear from across a room  with speakers rated for 87dB/W/M, because the distortion voltage alone gives 1 micro watt of power, which gives an SPL that is 60dB below the 4 watt level of 93dB, so the distortion produces an SPL at 33 dB which would be below the sound level of heartbeats, breathing, and natural background sound levels.

The THD spectral voltages in tube amps such as these at low levels up to 4 watts is usually a mixture of predominantly 2H and 3H with other 4H, 5H, 6H, 7H, 8H, 9H etc all at least 12dB below the combined levels of 2H and 3H indicated on the above graphs.
Either 2H or 3H harmonic may be greater at low levels because there is much variation in 2H at low levels due to unavoidable unbalance in the 2H currents in each half of the PP circuit because the 2H does not ever completely cancel as a result of push pull action because of slight tolerance differences in the matched tubes used in the output stage and driver stages.
Plus the input stage produces some 2H since it is a single ended triode stage.
The 2H content is the main reason for the differences between the two amps' THD. But at high output levels the THD of the pair of amps becomes nearly equal, and mainly 3H. The 2H at low levels can be minimized by placing tubes in either side of the PP circuit so that ac signal currents measured across the 10 ohm cathode resistors for each side have equal totals, or as close as one can get, and this is a very tedious thing to do and to monitor with a distortion meter.
Unfortunately, this needs some technical expertise to achieve, and usually does not lead to any subjectively more pleasing music.

Triode operation was tested but very little differences with THD were recorded at the same low levels.
Changing from UL to triode operation means the total circuit gain is reduced about 4 db. Therefore the amount of applied global NFB is also reduced by the same amount. With less global FB one would expect output resistance and distortion to rise
but the triode connection itself gives a compensating reduction in THD and output resistance so the change from
UL to triode does not require any change to the global NFB network resistances and the output impedance and THD
will remain very similar in either UL or triode.

These amps were prepared for someone with QUAD ESL63 and the amount of NFB isn't high and could have been increased to levels used more commonly by other makers who might use say 20dB, which would reduce all the above THD figures by about 9dB, or to 1/3 of the figures mentioned. I felt there was no need since the output impedance and THD was low enough.
 
There is nobody I know who can tell the difference between triode and UL connected output stages if the power levels are well away from clipping and if the general total levels of NFB are similar as I suggested above.

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