Fig 1,


Fig 2,


8585 General Description.
The basic design of the push-pull 8585 was conceived in 1995, following several years of trials with
various PP topologies to get the most accurate, dynamic, and subjectively pleasing sound,
along with the best measurements possible with the least use of corrective circuitry
known as global NFB.
The fundamental output stage circuit design of the 8585 is based on the renowned Acoustical Connection
of beam tetrodes invented about 60 years ago and incorporated in Quad II amplifiers designed by Peter Walker and in the much later Quad 40 amps with a similar design by Andy Grove.
The quality of the 8585 output transformers surpasses anything built for Quad amplifiers.
 
The October 2006 version 8585 is an integrated amplifier, with 6 switched line level inputs,
balance control, line level SET preamp amp, and volume control.
 
Tube Choices
Incremental improvements since 1995 evolved the present fine sounding amplifier.

There are presently ( October 2006 ) 4 x KT90 output tubes per channel, but the circuit allows the use of all the main octal based output tubes, such as 6V6, 6L6, 5881, EL34, 6CA7, KT66, KT88, KT90, and 6550.
However, only the 6L6GC, 5881, KT66, KT88, and KT90 are able to be plugged in and biased to suit the existing circuit.
The use of  6V6, EL34 and 6CA7 are usable, but require minor alteration to the circuit to ensure the correct grid bias voltage these 3 tube types and for 6V6 the
anode and screen supply must be re-wired to provide an anode supply of no more than +350V
and a screen supply of about +300V.

Local output stage cathode feedback and B+ Regulation.
The output stage for each channel has 4 output tubes, two for each side of the push-pull circuit,
and these are configured to allow 12.5% of the anode to cathode signal voltage to be fed back
locally to the output tube cathodes from a tertiary winding on the OPT.
With a 5 ohm load, the 12.5% of local NFB equates to about 7 dB of local NFB.

The reason for applying two lots of negative feedback, one lot in the form of local cathode FB  in the output stage
and the other globally is because the local FB in the output stage has a very favourable effect on the spectral content
of distortion harmonics which are reduced to lower levels than triode or ultralinear connection
but without paying a penalty of requiring too high a drive voltage to the output stage tube grids.
Less global NFB is then required to make up a total of less than 20dB and it is thus more possible to achieve without
requiring such critical stabilizing techniques.
If the output tubes were connected in pure beam tetrode mode the harmonic spectra would remain more
complex and the amplifier would be more difficult to stabilise.
The anode supply is +500v using an 800 VA toroidal power transformer,
silicon diodes, and large value capacitors, and a choke filter.
The screen supply is +325v, and actively shunt regulated.
The output tubes have fixed grid bias voltage applied, once the level of cathode current is adjusted, see below. 

The output stage is driven by a "long tail pair" differential amp using EL84/6BQ5 strapped as triodes and supplied with DC via a centre tapped choke and with 8.2k from each end of the choke to the EL84 anodes.
(A choke is a winding of wire around special grain oriented silicon steel, ( GOSS ) and it has high resistance to signal currents, but allows direct current to flow easily).
The novel and little used choke loading method in all my PP amps reduces dynamic signal current change in the EL84 to about 1/3 the usual level with a pure resistance supplying dc to the tube and the loads seen by the EL84 are mainly only the pair of 100k bias resistors
to each side of the PP circuit. Thus the EL84 are operating with a load of about 25kohms,
well above what might otherwise be used in a power amp using such tubes so
there is a very low level of harmonic and intermodulation distortion, since the higher the load on triodes, measured in ohms, the lower the thd becomes.
In this case, the choke was found to reduce thd about 3 times, or 10 dB,
compared to conventional circuits using cheaper dc carrying resistance loads.

Global and Local NFB
The input stage of the power amp is a 12AU7 with both halves paralleled.
The dc supply to this input stage is via a pair of 100k resistors in parallel
which means that although the anode voltage change is up to about 7Vrms, there is
not a huge anode current change in the parallel triodes.
The 12AU7 is set up as a single ended triode and there is 12 dB of "global" NFB from the output transformer secondary applied to the cathode via a low resistance network.
At clipping the amp requires about 2Vrms applied to the 12AU7 input grid
and there is 1.5Vrms of fed back voltage applied at the cathode and the difference between
the input voltage and fed back voltage is 0.5Vrms and this is amplified to make up
to about 7Vrms to power the following Long Tail Pair driver stage ahead of the output stage.
This basic method of globally applied NFB has been used for 60 years at least,
and there is a total of 19dB of NFB applied consisting of 7dB of local output stage NFB
which reduces the output impedance of the output tubes to about 2 ohms,
then there 12dB of global NFB reduces the 2 ohms down to about 0.4 ohms
which results in a damping factor of greater than 10 for a 5 ohm load.
The distortions in pure beam tetrode connected tubes is reduced by the output stage NFB by about 12dB and then the global NFB further reduces the distortions by another 12 dB.

Without connections of NFB the amplifier would be useless because the output resistance
would be many times the speaker impedances and the distortions would be
both audible and objectionable.

Power Output
The output power is 84 watts into 5 ohms at less than 0.3% thd which is at the onset
of overload or clipping, and is class AB1 with approximately an initial 20 watts of pure class A.
There can be 57 watts watts into 8 ohms at less than 0.2% thd, class AB1, with
the first 40 watts being pure class A.
Refer to graph BELOW for power output vs load to examine the maximum power output
levels possible.

Maximum Vrms signal levels into loads with the 3 output terminals strapped together are as follows:-
33 ohms, 18W, 24.4Vrms, 34.5 peak volts,
15 ohms, 32W, 22Vrms, 31.1Vpk,
8 ohms, 57W, 21.3Vrms, 30.2Vpk,
5 ohms, 84W, 20.5Vrms, 28.9Vpk,
2 ohms, 112W, 15Vrms, 21.21vpk,

Three pairs of Quad ESL will have an impedance varying from
11ohms at 60Hz to 0.6ohms at 18kHz.

This may seem like an impossible speaker impedance to drive but
90% of the power in music is within the 60Hz to 3kHz band where
Z is between 11 and 3 ohms, and only a tiny amount of power
is needed to produce frequencies above 3 kHz, and since the 8585 has
a power ability equal to 4 Quad II amplifiers in parallel and considering that stacking
the Quads will increase their power sensitivity approximately 3dB, then the
8585 will not have no more difficulty drive 3 pairs of ESL57 than a single pair of Quad II amps
will have driving one pair of ESL57 which are regarded as a compatible match of amp and
speaker since both were designed in the same era by the brilliant Peter Walker.

Even with just a 0.6 ohm load there is 60 watts of power available from the 8585
which means that the output current ability without clipping is about 10Arms, or 14 amps peak.

THD
At any load above 3 ohms,
1 watt of power, thd < 0.02%,
4 watts, thd < 0.03%,
16 watts, thd < 0.1%.
80 watts, 5 ohms, thd < 0.3%,
See the Graphs BELOW with following notes for Harmonic Products to view levels of THD with 5 ohms.

Frequency Response.
The frequency response for 1 kHz and 5 ohms is from 14 Hz to 65 kHz, at 80 watts, limited by saturation of the OPT at LF, and bandwidth limiting at HF.
The response widens from about 5 Hz to 68 kHz at ordinary loud listening levels.
With a test load comprising and RC series network of 0.5ohms in series with 6 uF and with 5 ohms shunting the
RC network there is no peaking in the response.
Pure capacitance loads of any value between 6uF and 0.1 uF may be connected
across the output terminals with the HF response showing less than 1dB of peaking at 20kHz, and
not more than 6dB of peaking between 20kHz and 200kHz, so the amplifier is stable
with any value of C load.
Tests were done on C loads at low output voltage levels of 1Vrms output to ensure that the
diminishing impedance of C loads at HF did not cause the active protection circuit to activate
because of excessive dc anode current draw from the power supply.
For example a 2uF capacitor has 2.48 ohms of purely reactive impedance at 32kHz and
if the output voltage level was raised to equal that with 2.5 ohms of pure resistance at
clipping at 1 kHz, the amp will shut down within a couple of seconds. 

Speaker and Amplifier Output Impedance.
Any type of load is permissible, including dynamic, ribbon or ES speakers,
and the amp will drive any load above 3 ohms for an average power level of 5 watts,
which allows for peaks in the music to be 50 watts or more.
There are 3 pairs of output terminals arranged so there is 0.6 ohms series resistance
from each output to the common internal connection at the amp so that
if 3 pairs of Quad ESL are connected with each pair taken to each pair of output
terminals then each pair of speakers is fed via its own separate 0.6 ohm resistance.
The output source resistance in series with the ESL is thus the amplifier Rout
of 0.4 ohms plus the 0.6 ohms giving a total of 1 ohm which is recommended for Quad ESL57
in order to get a flat response to 20kHz.
The 8585 will thus mimic the action of the Quad II amp but give greater stability
and lower distortions.
For normal speakers where the lowest amplifier resistance may be desired, all three
active terminals to each channel may be strapped with a wire beneath the 3 active binding
posts, and thus Rout = 0.4 ohms + 0.2 ohms = 0.6 ohms which
gives a damping factor of 10 with a 6 ohms speaker.

Sensitivity
With the volume control turned to the middle 12 o'clock position, approximately
4.9Vrms is needed for clipping power of 80 watts into 5 ohms which will produce
an ear deafening level of approximately 104dB SPL using 3 pairs of stacked Quad ESL57
based on being able to obtain 86dB using 1 watt at one meter with one ESL speaker from one channel.
At 80 watts the amp makes 20Vrms into 5 ohms and it does not exceed the voltage rating for the speakers. 
The level with 1Vrms of input which can be expected from a CD player
will reduce the output voltage to 4Vrms and a power output of 3.2 watts per channel
which would give an SPL of 94dB with the 6 stacked Quad ESL.
The gain of the internal preamp has been reduced to only
2.6 times or 9dB between its input and output so that excessive gain will not be a problem, yet there will always be a high enough power ceiling, regardless of the speaker variety used.

The gain of the preamp was somewhat carefully chosen because there is no
ability to delete the preamp from the signal path.

In-built Preamp
The 12AU7 integrated preamp ahead of each power amp uses both halves of the twin triode  paralleled, and there is an active constant current source for the dc carrying load component
to reduce the THD to tiny amounts regardless of the input levels which can be up
to 20Vrms before the preamp clips.
Even when the output level of the preamp is at 10Vrms with its input at 3.8vrms, THD < 0.3%,
and at normal levels where a CD player produces 1vrms on very high level signals
the THD < 0.06%, and nearly all second harmonic and thus not in any way
able to destroy the musical fidelity.

See the graph of THD for 8585 input preamp which shows
the 2H, 3H, 4H, 5H and 6H as they rise above the noise floor
between 0.4Vrms output and 10Vrms output.

The preamp has a mild 12dB amount of shunt NFB between its anode output
and grid input to ensure channel gain remains constant and well balanced
and to reduce THD and noise and Rout.

The input selector switch is a 2 pole x 6 position silver plated wafer rotary switch
supplied by RS components, the balance control pot is a cermet type
from Farnell, and the gain control pot is a dual 50k stereo Alps "Black" carbon track pot which has been used in numerous quality amps for the last 30 years at least
and which is available at RS and Farnell Components.

Tube Layout
The four front tubes are 12AU7.
The next row of four tubes from the front are EL84 or 6BQ5, which are exactly the same type
of tube, but with different commonly used type numbers.
The rear eight tubes are the eight matched octal output tubes.

Fuse Replacement and

PUSH-PULL amp operation.

The available undistorted power power varies with load. For the 8585 the recommended loading
for the frequencies between 100Hz and 800Hz should average 3 ohms or more.

Fig 4.


The graph curve shows the power output at less than 1% THD for loads between 0 ohms and 33 ohms.

Harmonic Distortion products in the 8585.
There are 3 following graphs with comments regarding harmonic distortion production.

Fig 5.


Fig 5 shows the distortion components of 2H to 7H where relevant, ie, as they increase with output voltage levels.
There is considerable 2H produced in comparison to the 3H at below 4 watts which would cover the
average listening levels of 99% of the population. The amount is all less than 0.03%, and lower
than an SE amp, and utterly negligible.
Notice that the dreaded higher number 4H, 5H, 6H, and 7H harmonics do not appear significantly until the amp is being worked above about 37 watts, or 13.5Vrms of output, where they try to rise above the 0.005% level.

Fig 6.
 
Fig 6 shows the difference in harmonic products produced when the input preamp
has its output minimised by turning up the volume control to maximum.
This gave less 3H and more 2H especially at high levels, and Fig 5 & 6 show the effects of 2H cancellations
when the preamp is used to produce more voltage before the volume control attenuation.
I first thought perhaps the preamp 2H would add to the power amp 2H produced in the
V2 12AU7 but this appears not to be the case, so the 2H is otherwise being produced in the power amp
to what i expected.
The point is that the 2H is a small amount of 2H for any tube amp.
In PP amps the 3H is usually the most dominant but in this design of PP amp the 3H
is very low compared to many other designs using  such low bias levels.
Being able to obtain less than 0.014% of 3H at 4 watts is a good result for any tube power amp,
and regardless of the volume control setting.
In both Fig 5 & 6 the higher order harmonics are negligible at ordinary very loud levels.

Fig 7.

Fig 7 has 3 curves drawn above for THD for the 8585 with KT90 output tubes and biased lightly with Ia at only 33mA per tube. Curve A & B were plotted in 2006 after changing from the 2004 schematic for the input preamp.
Curve C is derived from the 2004 measurements for 4,6,8 ohms in Fig 10 below, and for a 5 ohm load
and with volume control at maximum to ensure the input preamp creates the least THD possible.

Curve A is for the 2006 above amp schematic and with the volume control set at the middle position.
This means that any incoming signal is amplified 2.6 times, then reduced in level by a factor of 0.156
times by the volume potentiometer.
The curve A was plotted by varying the input signal at 1 kHz  until the amplifier clipped at just over 20Vrms into 5 ohms.

Curve  B was plotted with the same increasing input signal but with the volume control set to the
maximum level so to reduce the amount of amplification by the V1 12AU7 preamp
and negate its probable effect on the thd measurements.

Notice that there is a difference between thd levels  of curve A and B and it is due largely to
cancellation effects of the second harmonic distortions produced in the preamp and power amp.

Curve C is taken from measurements made on the older schematic presented in 2004.
There is not a huge difference between any of the curves, but  there is a consistent
reduction of THD of at least 6dB between the 2004 and 2006 versions of the amp,
ie, between the curve C and A respectively.
This is mainly due to a reduction of 2H in the V1 12AU7 preamp by using a CCS load for the anodes
and due to 12dB of shunt NFB to reduce the 12AU7 gain.
The older 2004 preamp relied only upon about 4dB of current NFB from the unbypassed 1.5k cathode
R of the V1 12AU7.
At low levels the reduction in distortion due to a large increase in NFB does not
result in an exactly proportionate THD reduction when the preamp is tested with the power amp.

Fig 8.


Fig 8 shows the THD result for the V1 12AU7 triode line level preamp built into the 8585.
I measured the harmonics up to 10vrms output only because when testing high output resistance
circuits above 600 ohms my test gear has to use a high input impedance low distortion buffer between the device under test
and the analyser whose maximum input voltage is limited to 10Vrms.
There is no point in measuring  output voltages above 10Vrms because it is extremely
unlikely the input preamp would ever have to produce more than this voltage level.
It is of course capable of about 60Vrms of output.
As can be seen the THD at 2.6 Vrms of anode output which would be a high level from a CD player input signal,
THD = 0.04%, and mainly 2H with ALL other harmonics below the noise floor.

There are those who despise and discourage the use of NFB to reduce noise, output resistance
and distortion on the grounds that a small amount of it such as I have used around this preamp
will significantly raise the higher order harmonic products above the 2H to become a serious sonic pest.
There is NO evidence that they are correct. I am convinced that 3H or other harmonics
all at below the 0.001% level do not have any effect on the sound quality at all.

Tubes do have some magic though, and it isn't spoiled by the presence of excessive distortion!!

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The older 2004 schematic for one channel of the 8585 amp is included here as Fig 9 for reference.
Part numbers used in Fig 9 have no similarity to numbers used in Fig 1 and Fig 2 above.
Fig 9.

The thd measurements were taken  with Ia = 40mA at idle per each output tube.
Should the bias Ia be increased to 60mA, the thd into 4 ohms which is the worst would be reduced from 0.04%
at 4 watts to 0.03%. The best fidelity is gained by using loads above 4 ohms.
It is thought that this reduction of thd by biasing more heavily into class A would not be audible,
and not worth the cost of wearing out the tubes sooner, and dissipating an extra 76 watts of heat within the amplifier.
The 2006 version of the amp measures better than the 2004 curves indicate yet the idle bias current levels are
slightly lower than the 2004 levels.

Signals from each output tube cathode of L and R channels are fed to a pair of differential amps.
The K1, K2, and  K3, K4 signals are from of each side of the PP circuit. Considering K1, K2, Left channel,
the two cathode signals are applied through R1, D1, and  R2, D2 to then be applied through 4k7 R3
and to the base of Q1.
A 100uF cap filters out nearly all the ac signal from the cathodes, and only the dc signal is applied to the Q1/Q2 differential amplifiers.
The highest dc signal from K1/K2 is what determines the dcV at the Q1/Q2 bases.
When all bias currents are correctly adjusted in each output tube, the dc voltages at Q1/Q2 bases will be very close to equal but should one tube vary its current by more than about +/- 10mA, there will be a dc change to Q1/Q2 inputs, and this will be amplified to produce a difference in collector voltages, and thus turn on the led through D4 to D7 diodes.
This will tell an owner immediately if there is a bias problem.

Diodes D3, D9, Q14, D20 from each of the base inputs of all 4 bjts Q1 to Q4 are taken to a common rail
which detects the highest dcV being applied to any base input.
The voltage on this rail is applied to Q7 emitter follower buffer base through R22.
Should the output of Q7 emitter rise to cause more than 0.8Vdc at the SCR gate, it will latch on and
the protection relay will open to cut the HT on the power transformer.

Turning the amp off then on after 20 seconds will reset the amp, but if the problem causing the excess cathode current
re-appears, then the HT will be shut down again.
With the shut down, both leds are turned on through D8 and D19.

Loadline analysis for one pair push-pull KT90 with 12.5% CFB.
Fig 12.

This maze of lines and curves is not a determined attempt to confuse everyone!
But any novice will be challenged so I suggest those should read the basics in the
Radiotron Designer's Handbook, 4th Ed, 1955, but I have provided a step by step
explantation for this example below .

All the tube characteristic dynamic anode resistance ( Ra ) data curve lines for each and every value of Eg do not need to be displayed to find the limits of power produced by a pair of PP output tubes. The only Ra curve needing to be drawn is the Ra line for one tube where Eg1 = 0V.

This curve is nearly approximately the same for the curve shown for beam tetrode use with the value of screen voltage
chosen.
It is the curve through points OABC in this case, and is slightly to the right of the same line for beam tetrode operation shown on the data sheets for this tube.
The load lines and power outcomes shown here are are also very similar to what could be obtained
with KT90 used in ultralinear with about 38% screen taps on the OPT instead of the 12.5% cathode feedback used to substantially linearize the the output stage operation.

Plate dissipation limits of 50 watts for KT90 also does not need to be displayed because we are considering
operation of the KT90 at idle at less than half their Pda rating, and with ac operation the tubes are never likely
to ever go near the rated Pda.

Three sets of loadlines are drawn on the same graph for 3 different anode to anode loads of
13,200 ohms, 6,600 ohms, and  3,300 ohms.
If the OPT for the pair of tubes had an impedance ratio of  1,000 : 1, then the listed a-a loads
would mean that the secondary loads would be 13.2 ohms, 6.6 ohms and 3.3 ohms respectively.
For best fidelity where a huge maximum power level is likely then a high RLa-a is required,
so an impedance ratio of 1,500:1 would be appropriate so that a typical 2006 speaker load of 4 ohms
is reflected to the tubes as 6,000 ohms.

Drawing the loadlines to find out operating conditions as follows:-

Consider the 6,600 ohm a-a loading.

1. Draw a vertical line for Ea, ie, the dc voltage between anode and cathode in the example above, Ea = +470V.

2. Divide RLa-a by 4, so 6,600 / 4 = 1,650 ohms.

This is the load which ONE of the output tubes will see when the other one has gone into cut off
during class AB or B operation. It is what is called the class B load for the tube.

3. Find the maximum Ia if  one of the two tubes was turned on fully to become a complete short circuit.
So Ia max = Ea / class B RL ; in this case Ia max = 470 / 1,650 = 284 mA.
Plot this value on the vertical Ia axis, see point Y.

4. Draw a line between Y and Ea=470V.
Now see where the line Y-to-Ea cuts the curve of the anode curve for Eg1 = OV; in this case its at point B.

5. Drop a line from B to the Ea axis, and read off the Ea value, in this case its +70V, and this is Ea minimum,
and the point to which the anode swings negative from its idle value of Ea, +470V in this case.

6. Subtract Ea minimum from Ea quiescent to get the peak load voltage swing at one of the two anodes.
So Ea swing  =  470-70 = 400V peak.

To find the power that is produced by the two tubes we need to know the total voltage applied across
the primary of the OPT. The swing down on one anode is matched by a swing upwards on the other anode.
7.  Anode to anode swing = 2 x Ea swing of one anode and in this case = 2 x 400 = 800 peak volts.

8. Multiply peak swing a-a x 0.707 to convert to Vrms. 800 x 0.707 = 565.6 V.

Clipping occurs in class AB amps where the voltage swing is limited by the Eg1 = 0V curve,
since going beyond that means we are driving the output tubes into grid current, or class AB2,
which we are not attempting to do in this case.

9. Maximum power output at clipping = V RL squared / RL.
In this case PO = 565.6V x 565.6V / 6,600 ohms =  48 watts.

This is the class AB power maximum and  is the same maximum for any useful values of idle current
between class B and class A.

The two tubes will operate at first in class A before moving to class AB above a threshold determined by the idle currents.

10. Draw point Q on the Ea line for idle. In this case Iaq for one tube = 40mA, and Q is on the Ea = 470V vertical line.
Draw a horizontal line of 2 x Ia, right across the page and so here we have a line at 80mA.,
and it will intersect our load line for 1,650 ohms at point E which is at Ea = +200V.
This point is where the increase in anode current in one tube will in theory equal the reduction of current
in the other tube, and class A operation is defined where equal Ia change in each +ve and -ve direction occur.

Now while each tube works in class A, its load is 1/2 RLa-a, so in this case its 6,600 ohms / 2 = 3,300 ohms.
11. Calculate peak class A load current that might occur if the tube was shorted and the load was the class A load.
Ia class A peak I swing  = Ea / class A RL = 470V / 3,300 = 142 mA.

12. Add the quiescent Ia because the tube already has Ia = 40mA.
Ia swing max + Ia q = 142 + 40 = 182mA.
Plot this point on the vertical Ia axis. It will be at point K in this case.

13. Draw a line K to Q and on to the horizontal Ea axis where Ia = 0mA, so that you will get point H at Ea = 600V.
This line is the 3,300 ohm class A load line  for each of the push pull tubes.

14. The class A power is determined by class A swing from anode to anode.
Class A peak swing, one anode = Eaq - Ea min for class A,  = 470V - 200V = 270 peak volts.
Class A swing a-a = 2 x class A swing at one anode = 2 x 270V = 540V peak.
Convert to Vrms, so Pk V x 0.707 , so 540V x 0.707 = 381.8 Vrms.

The anode to anode load will always remain the same regardless of the class of working of the tubes.
The Class A  PO =  381.8 x 381.8 / 6,600ohms = 22 watts.

You can see that the line through points YEQH has a kink at point E, and the load seen by one tube in
a PP amp isn't constant load during each wave cycle above the class A power limit occurs when one of the pair of tubes
has zero current, ie, tube cut off occurs.

This change of load during each wave cycle at higher power levels above the class A power limit is the cause of a change of tube voltage gain and accompanied by switching action which all makes odd number harmonic products increase
more in class AB than in class A. Raising the idle current will raise the class A swing so increase the % of class A power in the total AB power, and keeping RL high will also place the % of class A power high, as can be seen in the lines
for where 1/4 Rl a-a = 3,300 ohms, or RL a-a = 13,200 ohms.
In actual practice the sharply kinked character of the load line for an individual tube in an AB amp
does not occur because the transition from class A load to class B load is a swayed curve, so the
main distortion product is mostly only 3H and a little 5H which is offset by increasing
transconductance ( gm ) for the tubes above the idle current.

The loads for each tube while in class A will be slightly different because each tube cannot ever be exactly matched
in gm to each other, and so one tube may see 6,800 ohms while the other sees 6,400 ohms as a result.
This will lead to slightly different amounts of 2H distortion currents in each output tube which then
cannot cancel each other fully, so PP amps do often produce some 2H in their output distortions but it is usually
very much less than that produced in single ended stages, and less than the total amount of mainly 3H which all
PP amps produce.
The 3H produced in class AB amps such as this one is the main harmonic product but it is at a much lower level
than an SE amp's 2H rated for the same power.
Triode operation in class AB is considered to be the best and most acceptable by many because of the gradual cut off of current in the tubes by the applied Eg1, voltage, so 3H is low in most class AB triode amps, with little 5H.
Pure tetrode AB amps have the worst PP amp character and are much improved by the CFB connection
as in this amp or by means of ultra-linear connection.

Note that the two KT90 in class AB1 produce as much class A as a pair of 6L6 in pure class A, but more than twice the power in AB1, and with their idle condition causing very low stress on the tube because it is less than 1/2 the Pda rating
for the KT90 of about 50 watts.

KT88 and 6550 also offer a similar advantage over the smaller tubes such as 6L6, 5881, 6CA7, EL34, KT66 etc,
and offer superlative sonic performance with most modern speakers rated at 90dB/W/M, although
to get the best sound the OPT must have a high ratio to account for the tendency of modern speakers to have
dips in their impedance character of down to 3 ohms.