LOAD MATCHING
4
BEAM TETRODE PUSH PULL
AMPLIFIERS.
This page contains
:-
Beam tetrode background,
*Fig
1. Loadline graph for PP beam tetrode 6550 with 4k a-a load.
Plotting
loadlines for PP tetrodes, 17 steps to find maximum class AB power,
class A power.
Heat dissipation considerations and measurement,
92 watt
Class AB power with Ea = 600V.
Biasing the output tubes,
Distortion,
Output resistance.
Using a higher RL such as
8ka-a,
Global NFB, its effect on output resistance,
Calculation of amount
of applied NFB and the output resistance with applied NFB.
* Fig 2. Graph of power out vs RL
.
Loading the PP beam tetrode output stage, OPT ratios.
Ultralinear and
other output tube configurations,
Driver amplifier comments.
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Beam Tetrode background.
Some people
believe in using push pull beam power tetrodes for hi-fi without hesitation.
Probably the most famous of them
are the 6L6, 6V6, KT66, 6550 and KT88. We
now have the Russian made KT90EH.
The beam tetrodes have very close cousins
called pentodes such as the EL34 and EL84 etc.
The first tetrodes from the
early 1930s had no beam forming plates and made them a trifle unstable
but
otherwise much more useful for RF amplification than triodes. Someone added a
third concentric grid structure into a low signal tetrode and we had a pentode
which overcame the weird operation of the pure tetrode at low Ea
voltages
caused by secondary anode emission.
Soon there were power
pentodes about the same size as a power triode, but capable of twice the triode
output power.
All these inventions were patented and nobody could make
pentodes without paying a royalty. One tube making company
wanting to avoid
the royalties invented the beam tetrode where the beam forming plates perform
the function
of the third pentode grid, the suppressor grid. A patent for the
beam tetrode was granted, and that other company was happy.
Beam pentodes now
form the majority of power output tubes used in tube amplifiers around the
world, with most of them being the 6L6 in guitar amplifiers. For every ten
guitar amps around, maybe there is one hi-fi amp, and in the hi-fi amps
you
will rarely ever see a 6L6 because it is disdained because of its guitar amp use
but you will see KT66, KT88, 6550, EL34,
EL84, etc.
Many hi-fi
fanatics moaned when pentodes and beam tetrodes were invented.
The tetrodes
had atrociously anode resistance, Ra, and had much worse distortions than any
nice triode like the 300B
that came out in 1928 which worked fine without any
loop NFB. Tetrode ( or pentode ) amplifiers were required to have loops of
negative feedback connected to reduce output resistance and distortions, and
that kind of circuit jiggery-pokery in 1935
wasn't easy for the techs to get
used to because it required wide bandwidth output transformers and a good
working knowledge of practical ways to avoid amplifiers from becoming unstable
and oscillating when FB networks were connected.
But tetrodes were
twice as efficient as triodes, and a loop of NFB was cheap, the tetrode voltage
gain was high, so they were easy to drive, so the beam tetrode was here to stay.
The 6L6 and its cousin with a top cap, the 807, helped win WW2 for the allies.
Millions of 6V6 were used in countless AM radios as the 3 watt output
amplifier. The sound was often pretty awful when you turned up the volume,
and many radios didn't bother with any NFB loop. People didn't mind much though
because the source signal was often lousy anyway. The sound of a shellac
record transmitted on AM with a 3 kHz audio bandwidth
and 5% THD was a cause
to get drunk. Certainly if Marlena Detriech was singing.
So here we are
in 2006 with triodes, pentodes and beam tetrodes. Marlena went North, and the
sound sources have
vastly improved, and the vast majority of people use
transistors. Some Audiophiles have grudgingly accepted that you could connect
the screen grid to the anode and have a nice triode if you really had to have a
triode because a 300B is quite expensive for what it is and what it does, and in
my page on load matching to SE triodes illustrates what can be achieved with the
6550, KT88 or KT90 in triode compared to a 300B.
I prefer using a cathode
feedback winding which is 12% to 20% of the total primary turns for the multi
grid power tubes to linearize them. This is about 8 to12dB of applied local NFB
in the output stage.
Quad II made this CFB method the best known simple
application of the idea with 10% of the primary turns in the form of a tertiary
winding with centre tap which is about 8 dB of NFB.
For hi-fi amps the
Ultralinear connection has been the most popular since 1954 because it is
a simple matter of having two
taps on the OPT primary winding, rather than
have a separate "tertiary" winding with a CT for the cathode
feedback.
McIntosh went further than Quad with CFB where the OPT had two
equal turn windings with CT for anodes and cathodes
so that the amount of CFB
was 50% which is about 16dB NFB and 50 watts could be safely had from a pair of
6L6
in class AB beam tetrode way back in 1949, and with measurements better
than everyone else since there was also
20dB of global NFB applied .
There isn't much stopping anyone from using beam tetrode connection with
global NFB if you have a decent OPT and
enough NFB.
Most guitar amp
makers never use much NFB and nearly always use ordinary beam tetrode or pentode
connection
because the musicians like the distortions of pure PP beam
tetrodes or pentodes.
It is worth doing a graph for loadline analysis for
a pair of beam tetrode 6550 in class AB1
and with Ea = 400V, Ia = 80mA per
tube, screen supply = 350V, Pda = 32 watts.
Fig 1.
To plot the load line for one tube and calculate the
outcome for 4k anode to anode :-
(1), Establish the desired working
point for each tube at Q.
Ea = 400V and Ia = 80mA, Eg = -20V, Eg2 = 350V, Pda
= 32 watts.
(2), Calculate class B load = RLa-a / 4.
Class B load = 4k / 4
= 1.0k.
(3), Calculate Ia for Ea across class B load.
Ia = 400 / 1.k =
400mA.
(4), Drop a vertical from Q to Ea axis to plot point D. Plot point A
which is on the Ia axis and for the Ia current
calculated in (3).
(5),
Draw a straight line from A to D. This is the 1.0k class B
loadline.
(6) Calculate current change for the class A load line =
RLa-a / 2 .
Ia = Ea / RL = 400 / 2,000 = 200mA.
(7), Add Ia at Q to
Ia from (6).
80mA + 200mA = 280mA.
(8), Plot point E on the Ia axis for
the Ia found in (7).
(9), Draw the straight line from E through Q and
on to G.
(10), Mark the the intersection of the two drawn lines as
point C.
(11), Drop a vertical from point C to the Ea axis and read off
the Ea voltage.
The Ea voltage is 235V.
(12), Calculate class A
power for one tube, Power = ( Ea at Q - Ea at C ) squared / ( 2 x RL )
Power
= ( 400 - 235 ) squared / ( 2 x 2,000 ) = 6.8 watts.
(13), Calculate class A
power for both tubes at class AB threshold, = 2 x class A power from one
tube.
Power = 2 x 6.8 = 13.6 watts.
(14), Plot point B where ABCD
intersects Ra line for Eg = 0V. Drop a vertical from B to EA axis and read off
the Ea
which is Ea minimum anode voltage swing.
Ea = 65V.
(15),
Calculate maximum peak anode voltage swing = Ea at Q - Ea minimum.
Peak swing
= 400 - 65 = 335V pk.
(16), Calculate maximum anode to anode swing
voltage = 2 x pk swing at each anode x 0.707 Vrms.
Vrms a-a = 2 x 335 x
0.707 = 474Vrms.
(17). Calculate maximum class AB power = max Vrms
squared / RL a-a.
Power max = 474 x 474 / 4,000 = 56 watts.
Heat dissipation at the anode.
The input
power to the anode circuit when 56 watts of class AB power is produced will be
more than at the idle condition for both tubes when the anode input power = 2 x
32 watts = 64 watts. There will be separate screen input power which should be
less than 5% of anode input power.
The input power can be calculated for
class AB but I leave that to those of you who are able to follow the steps
in
the Radiotron Designer's Handbook. Its safe to assume that input power may be
double the idle condition.
The graph also has a Pda curve showing the maximum
limit for Pda = 42 watts. Strictly speaking the
class B loadline
should not cross over to be above the Pda limit
line. Where this occurs there is a danger than the power as heat
liberated
at the anode and equal to combined dc and ac x Ea in watts will
exceed the pda limit. When checking an amp after completion,
at maximum sine
wave class AB power, the power going into the CT of the OPT can be measured and
recorded.
Power dissipated in two tubes =
B+ power input - audio power output.
B+ input current can be
measured with a 10 ohm resistance between the OPT centre tap and the B+
supply
and the input power = total anode input current x Ea, dc voltage
between OPT centre tap and the cathodes.
Maximum allowable long term anode
dissipation limit limit is 42 watts for 6550, but short term dissipation of 60
watts for
a few seconds won't harm the tube.
In this case if the Pda for both tubes
using a sine wave test is less than 84 watts, all will be well, but even if it
is 20% greater, it won't matter with music since the average output power level
with music is well below the clipping level using a sine wave which is the most
arduous of all conditions.
70 watts of maximum output audio power could be
had with RL as low as 3k if the the amp could be driven into class
AB2
but its easier to simply use a higher Ea and remain in class AB1 so
a higher Ea swing can be achieved.
With
PP 6550 beam tetrode, the Ea could be 600V, Eg2 = 350V, Eg1 = -30V, Ia =
35mA, and RL a-a = 6k.
This would give 92 watts of class AB1 at about 5% THD
but quite low pure class A power of only about 4 watts.
Regulated power supply.
Such a class AB amp
will potentially draw much more anode dc from the B+ supply at full power so the
B+ must be fairly
well regulated. In the 1950s a tube rectifier and choke
input filter may have been used with a swinging choke of perhaps 20H and a
capacitor of 47uF.
But today any well rated power transformer with silicon
diodes will be better self regulating than the tube rectifier and choke input
filter, and the Si diodes allow a simple CRC or CLC where C = 235 uF or greater
and L = 2H.
Biasing the class AB PP stage should be done
using fixed bias because with cathode bias there is a big rise in the
Ek when
high level signals are sustained into class AB. But where the amount of class
power is equal to or greater than
half the total class AB power, cathode
bias is fine with music signals. Then the B+ supply needs to be higher to
accommodate
the extra cathode to 0V bias voltage.
Distortion in such class AB triode amps
will reach up to 5% at clipping depending on the amount of class A.
6550 in
pure class A beam tetrode PP can make about 35 watts in class A with 2% thd.
This is the best result in beam tetrode available, but if class AB operation
.
THD remains low until the AB threshold when the THD suddenly begins to
increase, however such class AB beam tetrode
amps are forgivable since the
the first 10 watts has a low THD level with mainly 3H with other odd orders well
below.
Output resistance is high
for all class A or AB beam tetrode amplifiers, and whatever load is used the
RLa-a will always be
much lower than the Ra-a. There will be a big change in
tube gain as the amp passes from class A to AB since
tetrode gain = gm x RL
and RL for class B is 1/2 the class A load.
The class A Rout where power < 2 watts = 2 x Ra-a /
OPT Z ratio.
Rout is lowest for the first few watts of class A power
because both tubes are connected to the load
because neither cuts off during
the wave cycles.
During the part of the class AB cycle where one of the tubes
is in cut off the Rout is theoretically double the class A figure
but
the Ra curves show Ra is lower at higher Ia so the effect of increasing Rout as
a result of class AB action is
reduced somewhat. Its important to realize
Rout for any beam tetrode or pentode amp is always going to be
much higher
than RL.
Assume *nominal*
speaker = 8 ohms, so use a 6 ohm secondary winding. OPT ZR = 4,000 / 6 = 667
:1.
Rout = 2 x 19,000 / 667 = 57 ohms.
The 4k : 6 ohms will also
give load matches 2.67k : 4 , not recommended for hi-fi, and 5.33k : 8.
So an
8 ohm speaker will be fine in a guitar amp using such loading. But for hi-fi, a
higher OPT ratio would be used.
Using a
higher RLa-a value.
The distortion and output resistance and amount of
class A of the above class AB1 PP tetrode amp can be
much improved by
raising the RLa-a to 8k by using an OPT with a ratio of 8k : 6 ohms for an
impedance ratio of 1,333 : 1 .
In the case above biasing of the output tubes
is best left at 80mA each.
Power will nearly all be class A and we get 35
watts maximum at about 1/2 the distortion or less.
At least 16dB of global NFB needs to be used to
reduce Rout to about 1 ohm.
THD will also fall to what may be acceptable
levels. 16dB of NFB will reduce 3% of THD to about 0.5%,
but that's at 55
watts, and at 2 watts THD without NFB might only be 0.5% so it will be
reduced
to around 0.1% by the NFB. Its still high compared to what class A
triodes with less NFB can achieve.
The
effect of NFB on output resistance should be explained similarly as in
the case of the single beam tetrode which I explain in my page on 'Load matching
to SE beam tetrode (1)'.
Consider the
case where we have an 8ka-a load, and we want to know what the Rout and
gain conditions are.
From the loadline drawn from point D for the the class B
of 2k for a tube, 9V peak grid swing is required
for full 375V of anode
swing into the class B 2k load for full power into 8ka-a AB1.
So Vin to each
6550 grid is 6.4Vrms.
Suppose we have a pair of 12AX7 in a differential pair
to drive the pair of 6550, and each has a gain of 64.
Since the 6550 grid to
grid voltage = 2 x 6.4Vrms = 12.8vrms, the grid to grid input voltage for the
12AX7
= 12.8 / 64 = 0.2Vrms for full power.
The anode to anode
voltage will be approximately 530Vrms, and since the turn ratio = sq.root of ZR
of 1,333 : 1,
= 36.5, the 6 ohm load voltage at 14.5Vrms.
Therefore we can
say that for 35 watts into 6 ohms we need 0.2Vrms input for 14.5Vrms without
NFB.
Open loop gain = 14.5 / 0.2 = 72.5.
If we have 1Vrms of NFB applied to one side of the
differential pair, the signal input will have to be 1.2vrms,
and the open loop gain
is reduced to 14.7 / 1.2 = 12.25 which is a gain reduction of about 5
times.
The
applied NFB = 20 log ( gain reduction ) = 20 x log 5 = 14
dBV.
The fraction of the output
fed back, ß, = 1Vrms / 14.7 = 0.068
Rout of the amp with FB applied
=
Ra-a
ZR x ( 1 + [ A" x {µ/TR} x ß ] )
Where Ra-a is the anode to anode resistance of the output tubes,
ZR is
the output transformer impedance ratio,
A" is the gain of the stages
preceding the output tubes,
µ is the amplification factor of the output
tubes,
TR is the primary to secondary turn ratio of the OPT = square root of
OPT ZR,
ß is the fraction of OPT secondary voltage fed back to be "in
series" with the input voltage to V1.
At the working point at Q Ra = 19k, so Ra-a = 38,000 ohms,
OPT impedance
ratio = 8k : 6 ohms = 1,333 : 1,
A" = 64,
µ = 190,
TR turn ratio
= sq.rt 1,333 = 36.5,
ß = 0.68
In this case, Rout =
38,000
= 1.2 ohms
1,333 x ( 1 + [ 64 x {190/36.5} x 0.068 ] )
If we have 2Vrms fed
back, then Vin = 2.2Vrms, so gain reduction = 2.2 / 0.2 = 11, so applied
FB = 21 dBV,
and ß = 2 / 14.5 = 0.137,
so the Rout
=
38,000
= 0.61
ohms
1,333 x ( 1 + [ 64 x { 190/36.5 } x 0.137 ] )
The previous samples are
approximate calculations of what to expect when you build and measure an
amp.
Notice that with an increase of applied NFB of approximately 6 dB,
there is a halving of Rout.
Fig
2.
The graph shows the power output for class AB1 6550 beam tetrodes in the
above load line analysis example.
This is a fair working point for the 6550
tube, because it offers the possible combination
of over 25 watts of class A
and a ceiling of 55 watts. People enjoy tube amps mainly because the first few
watts are so blameless. For the best musical performance, the first few watts
need to be class A, and without the sudden distortion
increase around the
transition from class A to AB known as crossover distortions. From the graph you
can see that if we
wanted the amp to produce at least 25 watts of class A,
then we would have to chose an anode load of about 8 kOhms, and we would have a
class AB ceiling of 35 watts.
Since most folks use no more than 4 peak watts
for 95% of what they listen to at home, the idle current could even be reduced
to 50 mA, and there would still be enough class A to cover all their needs,
without reverting to class AB, which would still provide the same ceiling of 35
watts, regardless of the tube bias current.
With the anode dissipation at 20
watts at idle, the tube life will be longer.
From the curves shown above, one can easily attain 50 watts of class A with a
quad of output tubes.
Speakers do not have an even impedance value, and often have high impedance
at bass frequencies, and lowest impedance between 200 Hz and 1 kHz, and some
have rising impedance above 20 kHz, due to the inductive nature of dynamic
speakers.
The load matching for the amp needs to be chosen carefully to
ensure the amp works within its class A ability for most of the time, even if
the speaker load value has dropped from the maker's nominal 8 ohms down to 4
ohms at say 300 Hz.
If the amp has an 8k : 6 ohm output transformer impedance ratio, then the
speaker loads from 4 to 8 ohms will be reflected back to the output tubes as
5.3k to 10.7k and we will always have just enough class A power to keep the
music sounding sweet, even with a 3 ohm speaker load. When the anode loading on
the tubes goes below 4 kOhms, all the distortions increase very
rapidly.
For the above working condition to
accommodate most modern speakers I would use an 8k : 6 ohm OPT.
All amplifiers have the same general shape for their graph of maximum output
power versus load value, including solid state types. Some makers make a large
fuss over the maximum power ability into some small value load but neglect to
say in their brochures that the same huge power mentioned is not available at
higher load values.
It should be obvious from the graphs here that if an amp
can produce 57 watts into 3.3 ohms, then only 18 watts are available into 16
ohms.
Every amp has a voltage swing limitation, and power
= voltage
squared
Load Resistance value in ohms
From the PP graphs, the higher the load value, the greater the % of class A
operation. This also gives lower output power, but lower distortion and Rout, so
load matching should be carefully considered.
Some tube amps have provision for connecting speakers
to different load matching terminals, usually 4 or 8 ohms.
When in doubt, always connect speakers to the lowest
impedance terminals.
The choice of impedance matching between 4 ohms or 8 ohms is determined by
the square of the turn ratio between primary and secondary windings on the OPT,
so if the turn ratio was 27.4 : 1 when using an 8 ohm speaker to the 8 ohm
output terminal, then the tube load will be 27.4 x 27.4 x 8 = 6 kOhms.
But
if there is a 4 ohm terminal, this will be connected to a tap on the secondary
winding,
and there will be effectively less turns on the secondary, giving a
turn ratio = 38.7 : 1,
and when a 4 ohm load is connected the tube load will
be 38.7 x 38.7 x 4 = 6 kOhms.
The tubes will thus act in the same manner
despite the difference in load values connected to the secondary.
Ultralinear and other output tube operation.
So far we have only
dealt with beam tetrode operation. But for many hi-fi amps the beam
tetrodes are connected
in ultralinear
with screen taken to taps on the OPT at between 30% to 60% of the anode windings
which overcomes the problem of controlling the high gain of the beam power
tetrodes when the load value is high.
The beam tetrode tube with 43% screen
taps have the characteristics of
Ra =
2.0k, µ = 15.4, gm = 7.7mA/V at Ea = 350V, Ia = 90mA, Pda = 31.5
watts.
The ultralinear, or UL connection reduces gain about 10dB
( 3 times ) for middle value loads but reduces the odd order distortions of the
output stage by about 10dB ( 3 time ) or more, and reduces the Ra from 19k to 2k
with a tap at 43%.
The characteritics of the UL connected single tube is in
my page on SE beam tetrodes.
The odd order distortion are reduced to near
triode levels in the single tube but the even numbered harmonics remain
When
the tubes are used in PP UL, the even order harmoics are cancelled by PP action,
so UL works to reduce the THD to being very close to triode but still
allow power levels near full beam tetrode connection and all it costs is 10dB of
gain so instead of 7Vrms to each grid for 8k anode to anode load,
we would
want about 22Vrms. This would mean that the diver amp would need to have two
stages such as a differential pair with 6CG7 and an input stage of 6CG7 because
not enough gain could be produced by a single 12AX7 differential pair
to
allow about 1.2Vrms input when 14dBV of applied global NFB was
used.
With 6550, about the same power vs RL characteristics as shown
on the Fig 2 graph for the beam tetrode example above is possible if Ea = 500V,
and with 50% UL taps. Eg2 will also be 500V because the screen taps are on the
anode primary winding which is at 500V. The increase in screen voltage makes Ia
higher, so Eg1 bias voltage must be increased to about -50V.
My page on '5050 integrated amp' has all the details about
UL connections.
The local cathode
feedback from the OPT is shown and explained in my schematics and notes for the
8585 amps I
have built.
But operating the tubes in ultralinear, you will get Rout of perhaps 6 ohms,
and the further reduction of the Rout to 1 ohm is achieved with some global FB,
usually about 16dB.
The use of 12.5% to 20% of the primary in the the cathode windings reduces
the Rout to as low as 2 ohms
with a further 8 dB of global NFB needed to
reduce Rout to less than 1 ohm.
The screens are capable of being used as
control grids, but as such, they do not have as high an amplification
factor, µ, so to alter the electron flow in the tube, a high signal voltage
needs to be applied to the screen to have much effect.
There are amps
designed to use a fixed grid voltage and have a drive voltage applied via the
screens so the output tubes then behave as a low µ triode with a very high Ra
similar to the beam tetrode value. Drive voltage needed is in the hundred volts
region and must be low impedance because the screen draw dc current at al times
which alters non linearly with applied
voltage. Screen drive is rarely ever
used even though it is sometimes more linear than conventional beam tetrode
operation.
In the UL amp, the screens do receive a high drive voltage from
the taps on the OPT, and it applied NFB.
When screens are connected to the
anodes, the NFB is the same as occurs in a triode.
In beam tetrode mode the screen is kept at fixed
voltage to prevent the the anode voltage signal change from affecting the
electron flow which occurs in triodes where the grid voltage changes AND the
anode voltage changes BOTH have a net mutual effect on the electron current
flow, due to electrostatic field effect action. The ultralinear method allows
the tube to operate
somewhere between tetrode and triode. With about 43% of
the anode signal voltage applied to the screen via the OPT taps on the primary,
the tetrode anode resistance will be reduced
from 17 kOhms to about 2-3
kOhms, and the distortions reduced substantially, whilst at the same
time,
the maximum output power is reduced by only 15% for the same Ea as the beam
tetrode case. With any more than 50% tappings, there are no more benefits to be
gained if we wish to keep the output power near the beam tetrode
levels;
going over 50% results in reduced power capability towards that
of triode connected tubes when 100% of the anode voltage is fed to the
screens.
In the 8585 Integrated amp I have used the distributed load connection, or
local cathode feedback windings on the OPT so the control of gain, output
impedance and distortions is localised in the grid circuit, as well as the
screen circuit.
This method is very effective in reducing the effective
anode resistance and distortion, yet maintaining the same power of the beam
tetrode. The other major benefit is that the screens can be connected to a fixed
voltage of just the right value to suit the output tubes best operating
performance setting for lowest distortion.
I used to think there is no absolutely best single way to arrange the tubes
in the PP output stage.
But I have not heard anything better than using
cathode feedback well implemented.
( Quad II uses 10% of the anode windings
for cathode FB to the KT66, so that the effective Ra-a at the OPT sec is reduced
to about 7 ohms and allowing for the winding resistance loss of 17% for the 8
ohm load setting, and with the added global NFB, Ro is reduced to about 1 ohm. )
I now make CFB standard on all my PP amps, unless I make a triode PP stage
which I sometimes use.
If 6550 beam tetrodes are set up with a fixed screen voltage with CFB or not,
then regardless of gain the Miller input capacitance is low and the output tubes
are very easily driven at high frequencies, so little drive current is required
in the driver amp. Having said that, I find that the best subjective sound
quality is achieved with a driver stage with a low output resistance
and
much more current and voltage swing ability than is actually needed than
calculations would predict, so in fact the
only time I would use a pair of
12AX7 as a differential drive/input stage would be for a guitar amp when we want
the sound to
be rather rosy and "flompy".
I prefer the CFB option, and when using about 20% of the primary for
CFB, the drive voltage to an output grid is raised to about 75 Vrms, or about 7
times the pure beam tetrode drive voltage so then the driver amp needs to have
more idle current and low distortion ability.
But I think the end result
sounds better with the CFB and mild global NFB.
With all the modes of operation including triode, when even a small amount of
global NFB is used, stability will depend on the quality of the output
transformer, and the design of the driver stage, and the control of phase shift
and gain at the extreme ends of the bandwidth of the amp.
Critical damping
is needed, and zobel networks to reduce HF gain and phase shift and
supply a
resistive load at HF to ensure stability.
The use of triode connected output tubes is another simple way to apply local
feedback in an output stage, but it comes with a penalty of reduced power output
capability, and higher Miller input capacitance, and a higher drive voltage of
about 35 vrms.
However, the use of a pair of 6550 in triode mode can produce
20 to 30 watts of excellent listening. If 25 watts is more than enough power for
you, triodes are the best sounding simple option.
If a slightly higher
ceiling is wanted, CFB is the next best, but much harder to implement because
OPTs with CFB windings wound properly with regard to winding geometry don't
grow on trees.
The problems of using evermore local NFB in the output stage become
counterproductive when the drive voltage required goes over 75Vrms to each
output grid, because the distortions in the drive amp begin to become serious,
so a balanced approach is required if we want to use a minimum of global
feedback, and still get low Rout,
and low distortion, and keep the drive amp
simple.
My experience with my 300 watt amps with a dozen 6550 tubes which now have
20% CFB applied in the output stage
indicates that I can still get
extraordinarily low thd with mild global NFB even though the drive voltages to
the output grids is 75Vrms to each grid. I use a pair of EL84 in triode and
these are choke loaded and in a differential amp.
The THD is lower than when
trying to 6CG7/6SN7 in a differential amp with resistances to supply the
dc.
Schematics and notes about the various techniques are shown in my other
amplifier pages.
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