Fig 1.
The above power
supply schematic was used for a 5050 integrated amp with two class AB1
channels.
Tubes used were 4 x 6550/KT88, with 4 x 6CG7 used for input and driver
tubes.
The above PSU
schematic suited the 5050 original only, produced in 2000,
The latest schematic has dual 6CG7 for each driver LTP, thus raising
the total number of 6CG7 wanted from 4 to 6
for the integrated amp.
Fig 2.
The design
process allows for the 2007 revised 5050 amp design, and includes
some safe margin for higher heater currents if different
input/driver tubes are used.
The explanation of
the design process includes a sample calculation,
and all steps in the design process are numbered :-
(2) Define the power requirements for
the
amplifier,
wanted secondary voltages and currents, and transformer VA rating.
The transformer must be able to easily provide
total output operating power needed for idle conditions,
and have a capability of sustaining 50% greater output power for a 50%
duty cycle
because of the change in input power to class AB output stages.
First
the Secondary winding VA outputs are all calculated, then summed, and
then primary windings calculated.
(2a) AC Heater Supply.
AC Heater supply for output and driver tubes,
4 x KT88, 6.3V x
1.8A x 4 = 45.4 watts.
4 x EL84, 6V6, or 4 x 6CG7 drivers, 6.3V x 0.8A x 4 = 20.16 watts.
Therefore the total ac power = 45.4
+ 20.16 = 65.56 watts.
One winding of 6.3Vrms must produce Iac = P / V = 65.56 / 6.3 =
10.4Arms.
Assume now that 4 parallel windings could be used, each having rating
of at least 2.6Arms.
Wanted windings = say 4 off 6.3Vac x 2.6A.
(2b) B+ Anode Supply for range of class A and AB
conditions.
KT88,KT90 etc,
Maximum voltage = 515Vdc,
maximum current = 60mAdc per tube at idle = 30.9watts x 4 = 124
watts at idle, but will increase because of class AB
increased power draw with music to about 200watts.
(The dc supplies will be measured from the
output of the first filter capacitor charged by silicon diodes).
For a voltage
doubler supply as shown, working Vac = Vdc / 2.6 = 515 / 2.6 = 198Vrms.
For a voltage
doubler supply as shown, working Iac, allowing for class AB Iac
variations = 2 x 124w/198V = 1.3Arms.
Driver and input stage B+ supply,
6 x 6CG7, 515Vdc x 35mA = 18 watts.
Allow for use of 2 x 6CG7, and 4 x EL84 or 6V6 etc, in triode for
drivers, 515Vdc x 100mA = 51 watts.
For a voltage
doubler supply as shown, working Vac = Vdc / 2.6 = 515 / 2.6 = 198Vrms.
For a voltage
doubler supply as shown, working Iac, no Iac variations = 51w/198V = 0.26Arms.
Note. The driver and input stages will derive power from the same 198V winding.
To this is added
power consumed by R across series electrolytics, and
any possible use of
shunt regulation at input stages, say 15 watts, so ac current from 198V
winding = 15w / 198Vrms = 0.076Arms.
Total B+ current from 198V winding = 1.3A +
0.26A + 0.076A = 1.636A
(2d) DC Heater Supply.
Allow generous power for two input
tubes, and attached external phono preamp say +16V at 1.6 amps, =
25.6
watts.
Other +/- 16V supply to
operate active protection
to switch open 198V B+ winding, say 2
watts.
Total power for 16Vdc = 25.6w + 2w = 27.6watts,
Idc = 27.6w / 16V = 1.725Adc.
For any full wave
rectifier with Si diodes as shown above, the unloaded Vac across
the CT winding = 1.41 x ( Vdc + diode voltage drop = 0.7Vdc ) = 1.41 x
16.7 = 23.55Vrms.
Allow for voltage sag of 5% when loaded, so winding Vac = 24.75Vac.
For full wave rectifier Iac = Pdc /
Vac across whole winding =
27.6w / 24.75V = 0.825Aac.
Allow +15% for higher dissipation because of rectifier peak currents, Iac = 1.1Arms.
Note. This winding should be able
to be used for ac heating for a variety of tubes.
Wanted winding = 12.6Vrms-0V-12.6Vrms x
1.1Arms.
(2e) Summary of transformer
secondary windings needed.
AC Heaters, 4 x 6.3V
x 2.6A = 65.5VA.
B+ Anode supply winding,
198V x 1.7A, = 336VA.
B - Bias supply of
50V x 0.26A = 13VA.
DC Heaters, 12.6V-0-12.6V x 1.1A =
27.7VA.
Sub total of VA for all secondaries =
442.2VA.
There will be heat
losses in the wire and core which we allow to be 10%
of the power supplied by all secondaries = 44VA.
The primary windings must be rated for secondary power + 10% =
442.2 + 44 = 486.2VA .
(2f) Primaries.
Want two windings with 120Vrms and taps to allow international mains
voltages
with series or parallel winding connections.
Allowing for 240V series connection, Iac = 486.2w / 240V =
2.02Arms.
Want 2 x 120V x 2A = 480VA.
The rating for this transformer is 480VA.
(3) Selecting the core type
and
size
for the transformer.
For excellent long
term operation we want to build a transformer that will
* not heat up more than 10Cdegrees above ambient,
* remains silent especially where we have a rectifier connected,
* be able to sustain a fault condition for 4 hours or indefinitely for
50% higher than normal current,
for any winding, and 20%
higher than usual mains voltages.
Thus I like to use
a
core with a 33% higher VA rating than the maximum mains input VA as
calculated above.
Remember the VA rating is the product of input voltage x input current.
The VA rating is a conventional term used by engineers and VA is volts
x amps as a rating instead of using
watts, which is power,
which is also the product of Volts x Amps working on a load.
So in this case
the core rating would be 480 + 33% = 638VA.
There several
choices for core material :-
1, Grain Oriented Silicon Steel E&I laminations, known as "low loss
GOSS lams."
2, Non grain Oriented Silicon Steel, known as "medium loss NOSS".
3, C-cores, which are usually always made using GOSS sheets rolled in
spiral around a rectangular mandrel,
glued well together, and cut to make two C shapes which have polished
meeting surfaces.
4, Unicore material available from AEM in Sth Aust, also made with GOSS
sheet material. This form of core
is far more difficult for the DIY person to use successfully.
5, Toroidal cores. Toroidal cores are made by winding a circular spiral
coil of GOSS material.
It is almost impossible for the DIY hobbyist to successfully use
unless he has an expensive
special toroidal winding machine.
I will base my
design on only using GOSS E&I laminations or C-cores.
In an earlier
edition of this website I said it was worth having a VA rating for the
power tranny of twice the
calculated VA input but with GOSS core material this isn't necessary,
and the 33% overload margin will be fine.
With plain non oriented silicon steel Si
Fe laminations or NOSS,
which is low grade transformer steel, there is justification for the
extra stack height involved with having a core VA rating
of twice the
VA draw from the mains.
This means that the core will be slightly bigger than if it were
designed for VA input + 33%
but it will have more surface area to
radiate heat and be more reliable.
NOSS which is 1/2
the price of GOSS has a similar Si content to GOSS but has not had the
rolling and heat treatments to
improve the permeability, ie, the µ of GOSS.
NOSS material in E&I cores when fully interleaved will typically
have a maximum permeability, µ, of 3,000,
but the GOSS will have a µ of up to perhaps 17,000.
This means that where one would get a temperature rise of 15 degrees C
over ambient for a generously designed
power transformer with NOSS, the same sized core in GOSS with the same
winding will perhaps give
a T rise of less than 5 degrees C after 4 hours, mainly caused by the
copper
losses being far greater than the core losses.
Toroidal transformers are nearly all spirally wound coils of GOSS which
have an iron µ of perhaps up to
40,000, because there is no change in direction to the crystalline
grain
which happens in assembled E&I cores.
The grain direction affects the magnetic properties, and the more
crossings of grain and
joint gaps, the lower the effective µ of the iron.
For most
transformers in audio equipment the B should be kept below 0.9Tesla,
and not up at 1.3 Tesla
which is typical for industrial/PA applications where the noise and
heat losses may be of little concern.
The audio amp power transformer needs to work silently and stay cool,
and also be well enough naturally regulated.
The hi-fi audio gear E&I power transformer will thus be
larger and heavier than the mass market industrial grade transformer
designed by accountants with poor
appreciation of hi-fi gear requirements.
The most simple
well known old equation to determine core size for
mains transformers has been :-
Afe = sq.root VA / 4.4,
Where Afe = T x S
= center leg core
section area in square
inches,
VA = calculated VA rating,
4.4 = a constant for high loss hot running worst iron,
or for where GOSS is used for low losses and cool running.
My examples will not be using this Imperial measures based equation.
For metric
measurement use ,
Afe in sq.mm = 146 x sq.root
VA, equation (a)
Where 146 is the
constant.
The most reliable
formula for checking on core size can be taken from the formula I
discovered
in an Electronics World article from about 20 years ago :-
VA = 4.44 x B x F x
I x ff x Sf x L x H x T x S
equation (b)
1,000,000
Where :-
VA = volt amps, or watts rating of the completed transformer,
4.44 is a constant for all equations and minimum grades of iron.
B is the magnetic field strength in Tesla, and we should aim for B =
0.9 Tesla for quiet running,
F is the frequency of the mains,
I is the current density in the copper, and we will allow 3.0 amps per
square mm,
ff is the fill factor, ie, the fraction of copper area in the core
window, = about 0.25,
Sf is the stacking factor of the laminations, since there ia a coating
of insulation, so about 0.95,
L is the length of the core winding window,
H is the height of the core winding window,
T is the core tongue width,
S is the core stack height,
1,000,000 is a constant for all equations.
This equation suits all types of cores, including those which are not wasteless pattern E&I.
However the
exception is with toroidal cores for which the simpler equation is OK
because
the window sizes of L and H are not known.
We could simplify
the formula for a a square center leg section knowing that S = T, and L
= 3T/2, and H = T/2,
Then L x H x T x S becomes 3T/2 x T/2 x
T x T = 3 x T to the fourth / 4.
Since the 4.4 constant will remain unchanged, and B will always be
about 0.9Tesla, F = 50Hz, I = 3Amps/sq.mm, Sf = 0.25, and ff 0.95,
then 4.4 x B x F x I x Sf x ff = 141.
Substituting the last two considerations into the above equation we get
VA = 141 x 0.75 x
T x T x T x T / 1,000,000
So VA = 1.06
x T x T x T x T / 10,000.
Therefore T
= fourth root of 10,000 x VA / 1.06,
Or T = 10 x square root of ( square root of
VA ). equation (c)
Where All the dimensions used are in
millimetres, and the pattern is wasteless and center leg is square.
The fourth root of any number is easy to
get, just enter the number, and press 'square root' button
of the pocket calculator twice.
At this point I
need to say something about core heating as distinct from heating
caused by copper getting warm.
All transformers will warm up even when the mains is across the
primary and no loads are connected to
any secondary.
In mains transformers the core heating is mainly due to hysteresis
losses in the core.
There is a formula which relates the "cosine of the current phase angle
to
losses" but that's all too difficult for you
to remember, and all that is needed here is to focus on the practical,
and the above formula will give you a
fairly
cool transformer even with relatively
high iron losses if you have B < 0.9 Tesla.
The Radiotron Designer's Handbook, 4th Ed, page 234 gives some figures
to use to estimate core losses
for various E&I laminations from an old maker's product called
Silcor which range at follows:-
4% Si steel, :- 1.32 watts per Kg @ B = 1 Tesla,
2.34 watts per Kg @ 1.3Tesla.
1% Si steel, :- 2.97 watts per Kg @ B = 1
Tesla, 5.04 watts per Kg @ 1.3Tesla.
Some steels I have
used which I took from old transformers obviously have high iron losses
because the iron
had a low Si content and it was not rolled and heat treated much and
got hot after a few hours of use.
The best steels I now use which have more Si content and better rolling
and heat treatments have lower core loss figures than the best in the
above list.
So if you wind a
transformer to the above formula with the poorest of NOSS iron,
it will work quite OK but don't be surprised if the temperature rise is
25 degrees centigrade
above ambient so that on a day when the room temp is 25C, the
transformer will be at 50C, and too hot to keep a hand on.
Nothing except the tubes should be too hot to keep a hand on in any amp
you build.
Using the same amount of GOSS with the above formula won't give you a
smaller transformer
but it sure will be cooler, and in
the case of the laminations I now
use from Sankey Australia,
the temperature rise due to core
heating is quite negligible.
The normally cheapest available E&I
laminations all have E and I shapes cut from bulk
sheet steel so no waste is generated. The resulting material shape is
called "wasteless pattern E&I material".
The wasteless
pattern material has a fixed relationship between dimensions seen
when an I is placed close
alongside an E and viewed from above. The size of the two holes seen
have "window length", L,
and "window height" H.
The distance across the center leg between each hole is known as the
"tongue" distance, T.
The fourth
dimension we are to know with a core is the height of the "stack" of
E&I laminations, S.
In all wasteless
pattern cores of any T size, L = 1.5 x T, and H = 0.5 x T.
VA = constant x B x F x
I x ff x Sf x L x H x T x S
1,000,000
VA = 638
constant = 4.4 for poorest iron.
B = 0.9 Tesla,
F = 50Hz,
I = 3amps / sq.mm
ff = 0.25
Sf = 0.95
L = 76,
H = 25,
T = 51,
S is the core stack height, and unknown.
1,000,000 is a constant for all equations.
Inserting above quantities we
already know into the above equation, we get
683 = 4.4 x 0.9 x 50
x 3.0 x 0.25 x 0.95 x 76 x 25 x 51 x S
1,000,000
= 13.67 x S,
So, S
= 683 / 13.67 = 49.96 mm.
We would rationalize this figure
and buy a pre-made plastic bobbin to suit a 50 mm stack of iron.
But let's use equation (c), knowing we want a
square core center section.
T = 10 x square root of ( square root of VA ).
= 10 x square root
of ( square root of 638 ) = 10 x square root of 25.25 = 10 x 5.025 =
50.25mm.
This agrees almost
perfectly with the equations above.
If the Figure for T was between two easily available tongue sizes, say
55mm,
we would perhaps have to apply the formula (b) and solve for S in the
longer way.
As regards
Toroids, if the center leg area was 50mm x 50mm, or 2,500sq.mm,
it too would have a rating of at least 638VA, but the benefit would be
lighter weight.
For C-cores, when usually arranged as two O's close to each other with
the wire around the
two close legs, T = twice the build up of the strips of one C-core, and
S = strip width of
the wound sheet. The equation (b) should be used because the C-cores
are not always
an equivalent size to replace wasteless exactly, and usually have a
bigger winding window area
relative to center leg area.
Standard plastic
pre-formed bobbins available for 51mm tongue lams have a range of
standard S heights,
say 51mm, 62.5mm, 76mm, and 100mm, ( ie, 2, 2.5, 3, 4 inches.)
( If we did
want the core to have a 2 x VA input of 1,366 VA, then S = 1,366
/ 13.67 = 100mm,
and we would select a
bobbin to suit a 100mm stack.
This will cost only slightly more in material and slightly more labour
compared to the 683VA rated core. )
The number of turns can be reduced and wire size increased for a given
Bmax, if the stack is increased.
Its actually
easier to wind fewer turns of thicker wire.
(4) Calculating turns per volt, TPV.
Notice I have
chosen the B = 0.9 Tesla, since this gives quiet performance with a
rectifier, although it does lead to slightly higher winding losses,
which would be slightly less if the B was 1.2 Tesla.
One would find a given transformer might be very quiet and hum free
when used to provide power to a purely resistive loading,
even when the Bmax is up around
1.2T.
However, if a rectifier is used to convert the transformer AC secondary
energy to DC and with a resevoir capacitance,
then the switching on and off of
rectifiers causes noise in the transformer as core material and
windings are jerked
around by the
pulsing magnetic forces.
Transformers are like electric motors, except that the windings are
restrained from moving,
quite unlike the movable armature, and nevertheless there can still be
some small amount of movement due to big physical forces in the
windings,
and this can be sometimes heard as hum in poorly made transformers.
An E&I transformer when mounted in an audio amp with full loading
should be quite inaudible until
one's ears are within 450mm away when ambient noise is low, say late at
night.
Many transformers in generic amplifiers fail this simple test.
Let me proceed based on selection of GOSS
iron with S = 50mm, T = 51mm.
This should run as cool as a cucumber with B = 0.9Tesla.
We need to know
the turns per volt for all windings. The TPV is the same for each
winding.
The next important universal transformer equation is
B = 22.55 x V x
10,000
equation (d)
F x N x T x S
Note that T x S is
the cross sectional area of the center part of the core which passes
through the winding.
We want now to
find the primary turns required.
The turns needed are not dependant on VA or current, but on voltage
and other equation factors.
In this
case,
using equation (f)
B = 0.9Tesla,
V = 240Vrms,
F = 50Hz,
T = 51,
S = 50,
N = primary turns.
Substituting,
0.9 Tesla
= 22.55 x
240 x 10,000 = 424.5
50 x N x 51 x
50
N
So
N = 424.5
= 472 turns for the 240Vrms mains, so turns per volt, TPV, = 472
/ 240 = 1.966.
0.9
So we would try to use TPV = 2.0 turns
per volt, or 480 primary turns. This will be subject to further
rationalization when we try to work out the wire size to fit nicely
across the bobbin width
without having to use an awkward fraction of a layer of wire anywhere.
Where possible, all windings should consist
of whole layers of wire of the same dia wire for all the main windings,
so layers of insulation are flat
across the wound layers.
(5) Check for iron heat losses in the
core.
Since we have
decided to use GOSS laminations, and the Bmax is 0.9 Tesla, nowhere
near saturation at over 1.4 Tesla,
we can work out the iron losses by
calculating the weight of the core and multiplying by the losses per Kg
of the material at the B used for the transformer.
The heat losses in watts per Kg ( or watts per pound ) are usually
stated by a manufacturer in their data for the core
and for a stated field strength, usually 1.2Tesla.
Volume of core for wasteless pattern
= ( 6 x T squared x height S ) / 1,000
where V = cu. centimetres,
T is tongue width in mm,
S is stack height in mm.
For our 638VA
transformer,
Core volume = ( 6 x
50 x 51 x 51 ) / 1,000 = 765 cu. cms.
Density of GOSS =
7.6 grams per cubic centimetre, so weight of core = 5,814 grams
= 5.81Kg.
From my private
data file on Sankey Laminations,
they list four types of steel lamination sheet used for iron
cored inductors and transformers,
and list them as follows, with the first 3 being non grain oriented
cheapest grade steel.
| Product name. |
Watts per Kg, 1.0 Tesla |
Watts per Kg, 1.5 Tesla |
| Lycore 150 |
1.5 |
3.6 |
| Lycore 230 |
2.3 |
5.3 |
| Lycore 350 |
3.5 |
7.7 |
| 35M5 ( GOSS ) |
not given, is low |
0.97 typical |
Losses per Kg at
0.9T are less than 1 watt per Kg, so for our 638VA transformer,
Losses in watts = 5.81 x 1 = 5.81 watts or less, which is only 0.91% of the input VA of the transformer.
If we had poorer
grade iron, with 5 watts per kg, the core loss would be 24.5 watts and
6%
of the input VA and along with the copper losses the transformer would
warm up considerably.
For those not able
to enjoys the splendid products from Sankey, the figures for M6 GOSS
laminations
in the USA etc will be about the same for the 35M5 Sankey material.
(6) Magnetizing current, iron µ check.
The
primary inductance can be calculated if one wants to predict what the
magnetizing current flow is in the core with no secondary loading,
or the µ of a sample of iron can be calculated if inductance is
measured.
Lp = 1.26 x N x N x
T x S x µ
1,000,000,000 x ML
Where
Lp = primary inductance in Henrys,
1.26 = a constant for all equations,
N = primary turns,
T = the tongue width,
S = stack height,
µ = the "mu" of the iron, sometimes specified, but also easily
measured.
1,000,000,000 = constant for all equations,
ML = magnetic path length in mm.
µ is
permeability, and is the number of times a given magnetic core increases
the magnetic field strength compared to the given winding wound without
any core.
An air cored inductor has µ = 1.0
µ varies
with the frequency of operation, reducing as F increases
and varies with the applied voltage, independent of whatever current
flows.
The variations of µ to applied voltage and frequency are non
linear.
In the case of
GOSS Sankey material, for 0.9Tesla, and 50Hz, µ = approx = 17,000.
For the 638VA core
with 480 primary turns,
Lp = 1.26 x 480 x 480 x 51 x 50 x 17,000
1,000,000,000 x 280
= 45 Henrys.
At 50 Hz, this will be an impedance of
ZL = Lp x 6.28 x F,
Where
ZL = the impedance, or reactance of the primary inductance, in
ohms
6.28 = 2 x pye, a constant for all equations,
F = frequency of operation.
For the 638VA tranny, ZL = 45 x 6.28 x 50 = 14,130 ohms.
We have 240Vrms
applied to the primary, so magnetizing current flow
= 240 / 14.13k = 17 mA, a tiny flow, and much better and lower than if
we used lower grade Lycore products.
To find out what
the µ is for the iron for a given applied mains voltage at 50Hz,
a small separate winding of say 200 turns is made on a temporary
bobbin to allow
a 12mm stack of sample lamination material inserted into the winding
with maximal interleaving.
Taping the laminations together with masking tape is sufficient for the
test.
Using a resistance in series with the coil of 1,000 ohms, voltage is
applied from a variac from the mains
through the coil and series resistance while monitoring the distortion
currents on an oscilliscope.
One begins the test setting the variac low at say 2vrms, and voltage
across the coil and across the 1,000 ohms is measured and recorded.
The voltages are recorded at 4V, 6V, 8V, 12V, 16V and so on until
distortion current observed just exceeds an easily visible 20%,
Armed with a set of listed VR, and VL, the impedance of the coil can be
worked out for each pair of voltages,
and a graph drawn of the impedance of the coil.
ZL = VL / IL.
IL = IR = VR / 1,000.
So ZL = 1,000 x VL / VR
With GOSS, you
should plot an arched shaped graph of ZL on the vertical axis and
applied voltage on the horizontal axis.
ZL will begin at 00.00, and rise rapidly, and then plateau, and begin
to fall, as the iron begins to saturate.
From the test
graph you can work out where ZL is at a maximum, and where inductance
is at a maximum,
and from the inductance you can find the µ because you have all
other quantities for insertion into the
inductance equation.
You can also work
out Field strength B in Tesla because you have applied voltage, Afe,
frequency and applied voltage and a
known number of turns.
You should find that the Bmax is between 0.6Tesla and 0.9 Tesla at the
top of the arched graph where L is maximal,
and distortion currents are low.
If the test is done with low grade material the distortion currents
will appear to be far higher.
One can apply a
similar test to any inductor or choke, with or without an air gap or DC
flow in a similar manner
to unlock all the secrets of the inductor performance and qualities.
If some second
hand laminations have become available for use there is no problem with
their re-use
in newly made transformers or inductors providing the iron µ is
high enough to allow
the use application without generating clouds of smoke. Hence with
second hand iron,
the above testing is essential.
(7) Working out the winding layers
and turns
The aim of this section
is to design the winding layout,
and draw up a bobbin diagram plan for the workshop to enable winding
the
transformer
with precision, and without confusion.
The final assembled
transformer should have at least one labeled termination board for
winding ends
mounted on one side of the bobbin, or suited for potting, with the pot
and core mounted above
the board, with terminals facing down into a chassis space when mounted
in the amplifier.
It should be decided at
the outset how the transformer is to appear.
All exposed terminals must be
boxed or concealed for safety reasons!
The area of the
window is equally divided for Primary and Secondary windings.
The basic sequence of windings will be :-
(1) Mains Primaries,
(2) Output tube AC heater windings which form an electrostatic shield
between mains primary and other following
secondary windings.
(3) HT or B+ windings for the anode supply
(4) Bias windings,
(5) Other minor low voltage AC heater windings for AC or DC heater
supplies.
The current
density design rating in every winding will be be 3Amp/sq.mm with
tolerance +/- 10%.
Some windings will have less current/sq.mm because otherwise the wire
be too thin to be
easily wound,
and too fragile in the case of an overload.
Wire sizes
selected are measured in their diameters in mm, not gauges.
Where wire size is given as say "1.0mm wire", this is the copper
diameter, and not including the enamel thickness
making the overall dia including enamel = 1.093mm.
Wire is
polyester-imide enamel coated grade 2 magnetic
winding wire with two applied layers of the enamel, and a chart of wire
sizes is
given elsewhere on this website, formerly donated to me kindly by
Blackburn Wires in NSW, Australia.
Plastic
transformer bobbins for mains transformers are available with a
pre-moulded plastic vertical dividers
of 2mm thick to keep all mains primaries isolated from all secondaries.
Sheet insulation
should be polyester, known as Mylar, or Nomex insulation available from
wire and transformer
parts sellers in a range of thicknesses, 0.05mm, 0.1mm, 0.19mm, 0.25mm,
etc.
Also suitable for where thickness is above 0.25mm, flexible cellulose
electrical grade fibre board is suitable,
and has its durability raised when it is fully coated and penetrated
with varnish.
If we do not wish
to use a vertically divided bobbin, then there must be a 2mm thick
concentric insulation layer between primaries and secondaries, or as
stipulated
by the electrical codes in your country where you are winding your own,
or
as required by the country to which you are exporting.
The sample chosen
for design is the 480VA transformer as nominated
previously.
The usual plastic
insulation thickness allowance for mains transformer moulded bobbins on
each side flange"cheek"
and bottom base between core and windings is 2.0mm for core tongue
sizes between 38mm and 62.5mm.
Allow clearance
between
final low voltage winding and core = 1mm.
Insulation
thickness between mains primaries and and first on sec = 2mm.
Bobbin base thickness and clearance off core = 2mm.
Therefore total minimal
insulation and clearance = 2mm + 2mm + 1mm = 5mm.
Thus the maximum winding height
available in any bobbin for transformers between
300VA and 700VA including all
minor thickness layers of insulation = 0.8 x window height H, or H -
5mm.
The maximum winding traverse winding width = window length L - 4mm.
240Vac primary, 7
layers x 0.99
mm-------------------------------------------------------------- 6.93mm
Insulation = 4 x 0.05mm-
-------------------------------------------------------------------------
0.20mm
Insulation between halves of 2 x 120V mains
windings, 1 x 0.2mm -------------------------------- 0.20mm
Mains Insulation
between primary and following secondaries, 1 x 2.0
mm--------------------------
2.00mm
6.3Vac heater
windings, 1 x
1.279mm-------------------------------------------------------------
1.28mm
Insulation
between ac heater windings 0V to B+ at +260Vdc
max,---------------------------------
1.00mm
200Vac HT = 6 x 0.99
---------------------------------------------------------------------------
5.94mm
Insulation 5 x
0.05mm-----------------------------------------------------------------------------
0.25mm
Insulation between
HT at +256Vdc and following bias winding at
-70Vdc--------------------------
1.0mm
50Vac bias winding, 1 x up to 0.6mm
------------------------------------------------------------- 0.60mm
Insulation cover
over
bias
winding = 1 x 0.5mm ---------------------------------------------------
0.50mm
bobbin bottom
thickness---------------------------------------------------------------------------
2.00mm
tape over completed
winding-----------------------------------------------------------------------
0.20mm
___________
Sub total
------------------------------------------------------------------------------------------
22.10mm
height of
window-----------------------------------------------------------------------------------
25.00mm
Clearance between
whole wind up and iron
-------------------------------------------------------- 2.90mm
Overlaps in insulation.
Insulation is
overlapped a couple of centimetres but only on the
top and bottom
of the bobbin away from where overlaps could fatten the wind up to
prevent lamination insertions.
Keep track of the turns and layers!!
Turns must be
counted from the beginning and noted in the wind up diagram in the work
book
so that one knows where one is up to and one does not place an unwanted
extra layer or leave one layer out,
which negates all your work.
Taps.
Wires from taps
should consist of two wires brought out from where the tap number has
been reached,
and another wire taken in to begin continuing. These must not be
twisted, but must be generously
sleeved with woven material to avoid shorts at the crossed and exiting
tap wires.
The tap wires must be neatly arranged and dressed by hand, so that
consequent layers will neatly run over the tap wires
without trouble from local pressure points.
The tap wires mean windings must be held tight
while fiddling with loose ends by thin clear adhesive poly tape, and
all while sticky with varnish.
Label all taps and winding loose ends.
All winding taps and winding ends should be labeled with a piece of
masking tape and felt pen with numbers for mains voltages
and letters for secondaries, as indicated on the winding diagram.
Sleeve before soldering wires.
Before soldering
wires to terminals, use sleeving from back of terminal to bobbin wire
exit if wire sleeving applied
at in-out points isn't long enough to get to the terminal.
Fasten loose ends as you wind.
During wind up on
the lathe, the emerging ends of windings should be secured around
screws placed in
plywood plates clamping bobbins tight on their mandrel. Coordinate this
carefully, lest you end up in mess,
or yank an end of a winding and break a wire, which utterly negates
your work.
Potting.
Potting is an
additional process I sometimes use after varnishing to make a
transformer or choke mechanically quiet.
I make a sheet metal box with suitable 1mm thick sheet iron, and set up
the tranny ( or choke ) inside the pot so it is
bolted in but also with provision for bolting all to a chassis.
I use molten roof pitch heated up in a steel pot on a camp stove in the
yard and also pre-heat the pot with tranny to about 100C.
The molten pitch is just poured in until the can is full. The minimum
space between transformer and any part of the pot
should not be less than 6mm, to allow pitch to run in without
solidifying quickly and leaving a little air void.
Sometimes I do it in two hits to allow for some shrinkage with pitch
when it solidifies.
You will find the
pitch adheres remarkably well to all things in the pot, and it can be
melted out in future
if a re-wind is needed, by placing the pot on a hot plate, and pouring
pitch out.
Pitch has been used for potting for 100 years at least.
The terrible smell of hot pitch is like having the road repaired
outside your house,
except worse.
Caution, precision work, leave all
alone if it all seems too hard.
For those not used
to detailed precision craft work which takes enormous amounts of time,
and takes several learning attempts, then they should never attempt to
wind any transformer.
Plenty are available from Hammond Engineering are quite suitable.
Winding losses.
I could
calculate the winding losses, but when all windings are rated for close
to 3A/sq.mm, there is no need because
winding losses will always be OK.
Copper losses for each winding = DC
resistance x I squared.
I suggest I leave
that for you
to check out, and if you come up with a better lower loss transformer,
itemize your findings to argue your case with me.