The input tube is a 6CG7 twin triode with both halves in parallel.
The parallel connection gives a single triode with Ra = 5k, and µ = 20.
Its DC supply is via the MJE350 wired as a constant current source with a very high finite resistance of many megohms. Transistors wired like this have no negative effect on the sound, and increases the fidelity because the constant current source
acts as a passive high impedance current supply. The triodes do all the actual work on the signal amplitude.
The input tube distortion of this stage is reduced about 10dB compared to using a resistance to deliver dc to the tube
because any such resistance becomes part of the ac signal load, and the higher the value of anode load ohms,
the more linear a triode becomes. The 6CG7 is electronically the same as a 6SN7, and the data curves for
either tube are the same.
The input triode has its own shunt regulated anode supply voltage. Its DC heater supply is derived from rectifying the AC heater supply. This input tube can provide wide enough bandwidth to power the long tail pair using a pair of EL84 in triode mode.

Alternative input tubes could be used for an input tube, but the heater supply ( sheet 8 ) may have to be changed to suit, since twin triodes such as 12AT7, 12AU7, 6DJ8 and others have different heater pin layouts and current
or voltage requirements.
Just about all other tubes could be used with the CCS anode supply as shown but the cathode R4 would have to be changed so that the correct anode dc voltage appears. Where anyone might use an input tube with different gain, the global NFB network R21, R22, C8 will have to be changed.

I doubt there is a better input tube than the 6CG7, ( or 6SN7 if people prefer octal tubes. )
My tip for the week is that Sieman's NOS 6CG7 are *very* good sounding, if you can ever buy any without having to sell your house to raise the funds.

The long tail pair, ( LTP ) or otherwise known as a differential amplifier has two EL84 wired as triodes because these small power pentode output tubes equal the performance of five paralleled halves of a 6CG7, but have slightly better linearity.
This LTP stage acts to convert the single ended unbalanced output of V1 into two exactly equal output voltages of opposite phase to drive the output circuit. About 75 Vrms is needed to drive each output tube grid at clipping, and about 8 Vrms is needed at the one live input grid of one of the EL84s.
Both input grids are biased from the voltage established at C5.
Note that C7 and R11 form a low frequency step response in the open loop character. This helps stabilize the amp at bass frequencies and improves recovery on overload, reduces saturation effects, and improves the margin of stability at LF.
Bass sounds tight and well controlled.
There is no HF step response network shown because it was found that this amp didn't need such a network to have flawless unconditional stability at HF, but some zobel networks in the output stage did reduce the inevitable slight square wave overshoot into purely capacitive loads..

This LTP is unique in that there is a choke of 500 Henrys with a CT in the anode circuit as well as the resistor loads of 6.9k to each anode circuit, R17,R18, and R19,R20. This combination of choke plus resistive load provides a low dc supply impedance which has very high ac impedance, and the main load each of the EL84 tubes work into is the capacitance coupled combined
6 x 120k ohm grid bias resistors of each side of the the PP output stage.
All the bias supply ends of the 120k resistors are bootstrapped to the cathode feedback winding through Cc.
Thus the load see by each anode of the the LTP is approximately 40kohms, low enogh to ensure good ac balance
but high enough to ensure THD of the driver LTP stage is about 10 dB lower than if the dc was brought to the EL84 anodes via resistors that would have to be approximately 15k, which is really too low to get low THD.

The 6.9k ( R17+R18 and R19+R20 ) loads isolate the shunt C and shunt L of the CT choke from causing much phase
shift at HF and LF. The very slight loss of gain at extreme ends of the band, ie, below 10Hz and above 30 kHz is
beneficial to stability with NFB.
The Ra of each EL84 with 15mA of DC is only 2.2 kOhms, so the drive amp has a wide bandwidth when driving 6 output tubes. The input Miller capacitance of the output stage is very low because a fixed bias voltage is applied to all the output tube screens.

A transistor constant current source is used to feed the commoned cathodes of the EL84. This type MJE340 transistor acts
as a purely passive manner, and has no discernible effect on the sound, and acts to maintain the balance of LTP output voltages within 1%.
A  pentode tube could have been used instead, but there would have been no benefits.
In an earlier version of this LTP, some local unbypassed Rk were tried in both EL84  which halved the gain but it was felt this robbed some dynamics because the effective Ra of the EL84 is made greater by adding  µ x Rk to the Ra without the Rk.  
In this circuit with cathode feedback in the output stage the output tube grid signals are twice the levels of the earlier ultralinear configured amps so if the gain of the driver is only about 9 with local Rk current FB, then the signal from V1 becomes too large to get low distortion, so LTP needs to be set up with maximum gain, to keep the output from V1 at a minimum.

I have found dc condition drift is negligible over many years since the EL84 are set up with a very low amount of Ia
compared to when used in an output stage where they usually run at Ia = 40mA but here have Ia = 15mA.
So the EL84 should last a long time without problems.
The THD from such an LTP is mainly 3H, but below1% even at 100 Vrms from each anode, and in this circuit a maximum of
only 75Vrms is required Some 2H is generated if the two tubes are not matched, and to minimize 2H
in an average pair they can be reverse positioned to get the best 2H cancellation between V1 and themselves.

At normal levels and because of the global NFB the THD is quite negligible since THD is propoertional to output voltage.

A previous version of this driver had zener voltage regulation for the bias voltages for the
CCS base and the grid voltages, but that was found to be unnecessary because these voltages and that of the V1 triode
is supply from a zener regulated supply of +320V, see sheet 4.

There some obviously acceptable other tubes that could be used in this circuit.
EL86 are pentodes which will work with their B+ supply to the choke at  10% lower than shown because
their Ra is only 1.4k in triode and you get a slightly wider Vswing at the anode.
However, their gain is only 1/2 that of EL84 since µ = 11 only, so when producing 75Vrms from each anode,
about 17Vrms is needed to drive the live input grid of the LTP.
The 6V6 could also be used and the gain is very similar to EL86, but the Ra of the trioded 6V6 is
higher than EL84, but nevertheless sufficiently low to provide excellent driving power to a multi-tube output  stage.

The all octal tube input stage would probably look better than what I have used; 6V6 and 6SN7
are plentiful and many NOS varieties exist, and like 6CG7 and EL84, the sound is just DREAMY.

OUTPUT STAGE FOR 300W AMPS, sheet 2.

The output stage sheet 2 above looks complex, but it is mostly just repetition of a basic idea.
The R&C part identification seems strange but all resistors and capos in similar functions for each tubeare just labled with the same number for R, and same letter for C; I am sure any tech will get used to the idea.

The two balanced outputs from the LTP driver amp is fed to two rails from which there are 6 coupling caps of 0.47 uF, Ca, each to couple each output tube.
2k2 grid stoppers, R1, are used on each tube to prevent RF oscillations.
Each output grid is biased with 120k, R2 taken to a -14V fixed bias supply, which is shown on sheet 8.
Each output tube has 15 watt cathode bias resistors comprising three 1.5k at 5 watts each, R3..
The cathode bias voltage will be about +18V, so total fixed plus cathode bias = 32V.
This method gives good self regulation of the bias and saves having so much heat dissipation in so many cathode resistances.

The anode supply is about +500V, and a fixed voltage is supplied to the screens via separate 330 ohm screen stoppers, R4.
At no time did I find that the tubes wanted to oscillate at some frequency well above the audio band.

The 6 cathodes on each side of the PP circuit with their RC bias networks Cb+R3 are taken to the ends of the cathode feedback windings but via 1.67 ohm current sensors, R5.
Voltages across T-U and V-W are used to work dynamic bias stabilizing circuits shown on sheet 5.
Capacitors of 470uF Cc are used to bootstrap the applied fixed bias of -14V so that the 6 x 120k bias resistances are seen by the driver LTP as values which are approximately 2-3 times higher than the actual 120k loads, so the actual
load the driver amp sees is about 40k-60k ohms on each side plus the shunting choke ( see sheet 1)  which is negligible since the choke impedance is so high.
This mild form of bootstrapping helps the low output impedance driver amp work with minimum distortion.

Resistors of 4.7k, R6, form negligible loadings on the CFB winding, but allows the bootstrapped -14V bias voltage to be supplied to the 6 x 120k bias R.

The Output Transformer has well interleaved windings and provides bandwidth at 250 watts at 15Hz to 270kHz.
Not all the HF bandwidth is used when global NFB is added, and bandwidth is reduced with NFB to 65kHz
for stability reasons.
The 12 secondary windings can be arranged to give waste free and uniform current densities
in all secondary windings to match from 1,200 ohms anode to anode to either 2.5 ohms or 5.6 ohms.

It is a somewhat complicated task for a non technical person to change output transformer matching, so the default setting is the 5.6 ohm load match.
The amp will still cope with 4 ohm loads since there is so much available power.
The tube load value was chosen so that it is equivalent to having 7,200 ohms per pair of output tubes.
This load could be as low as 4,000 ohms without causing undue increases in distortions or loss of power so when the amp is set for 2.5 ohms load match,  1.4 ohms could be driven.
When set for 5.6 ohms the load could be as low as 3 ohms without a problem.
Some speakers have DISGUSTING impedance curves with dips in Z well below their nominal
claimed Z but this amp design will handle them all with ease.
 
Maximum power output of approximately 350 watts would be when the load is about 3 ohms using the 5.6 ohm load match. When 8 ohms is used the amp gives 200 watts of output and with a very high % of class A power.
The dominant idea with this output stage is to produce a large amount of class A power but with a good reserve of class AB power for loud transients.
The function of the 300 watt output stage is no less than having six of my 50 watt amps all wired in parallel.

But should the voltage at one or more cathodes ever rise to levels resulting in a possible early tube failure, ie, from 35mA to say 64mA, then the cathode voltage will rise from about 18V to 32V, and the active protection circuit will turn off the anode supply. The protection circuits are shown on sheets 5,6 and 7.

The only stabilizing zobel network needed is the 4.7 ohms + 0.22uF across the output terminals.
Thus at 154k, ZCe = Z 0.22uF = R7 = 4.7 ohms so as frequency rises above 160kHz there is an
increasingly resistive load applied to the output terminals.

Any value of pure capacitance load across the output terminals does not provoke any HF oscillations.     

Back to Index Page