The input stage consists of a couple of general purpose small signal
transistors which are arranged as a longtail pair amp,
(differential pair
amp), with a low noise BC109 as a constant current source for the common emitter
circuit. The two halves of the LTP have some local current FB via the 47 ohm
emitter resistors, R11,12. The LTP is quite
linear in its operation, and only needs to generate up to about 1Vrms of signal
at each collector to produce full power at the output.
The output stage uses four N type Hitachi output mosfets which are
fed with a +35 volt 3 amp supply. There is a wide bandwidth output transformer
using C-cores, about the same size as a 50 watt OPT on a tube amp to couple the
mosfets to the load. The OPT has a ratio of 35:5 ohms, and has thicker wire than
a tube amp in the primary, but has
a 5P x 6S interleaving patteren to get a
300kHz HF cut off. The dual C-cores have a small air gap to reduce the effective
µ to about 2,200 to prevent saturation effects below 16Hz at full
power.
Since mosfets can generate much higher peak current than any tube, the
option of air gapping the PP core
resulted in much reduced current spikes
below 16Hz at full power. This opened my mind to using gapped cores
cores
for any PP transformer. With modern GOSS core material the µ of the core may be
much higher than it needs to be
and a reduction with an air gap may not
substantially reduce the primary inductance that shunts the RL and Ra-a
or
the RL and Rd-d, the drain to drain resistance of the mosfets. Modern GOSS I use
with E&I lams maximally
intermeshed will give a µ max = 17,000, and it
could be 1,700 without causing any loss of response
at bass frequencies
above 14 Hz.
Each mosfet has an idle current of about 0.7 amps, and dissipates 24
watts each, (like an EL34).
This is the reliable continuous limit for a TO3
package device imho. The drain to drain load is 35 ohms, so each half of the PP
circuit has a 17.5 ohm load in class A, and thus each mosfet sees 35 ohms of
load. Each mosfet is biased with 0.8 amps
of idle current, and there is
a peak current swing in each mosfet of +/- 0.8 amps peak. The voltage swing at
the drain is about +/- 20V peak, so each mosfet produces 11.2 watts, which makes
a total of 44.8 watts class A for the 4 mosfets.
The mosfets is biased in their negative temperature coeficient
region which means that if they heat up the bias current tends to reduce. Unlike
power transistors, they don't run away thermally due to the positive temp
coefficient.
Therefore the mosfets may have have truly fixed bias set by the
resistive divider of R21/R22, 32k and 1k, providing about +1volt bias to the
gates via the 68 k bias resistors. There is cap coupling from the driver stage,
but because there are only two stages in the is amp, the stability at LF is
excellent. At gross overload, the caps charge up like they do on a tube amp, and
the mosfets are driven into a temporary state of harmless cut off. This only
occurs when the power exceeds the
clipping level, and there is zero need to
worry about the effect under normal operation, just as is the case with a good
tube amp.
The 7.5 volt back-to-back zeners control the applied voltages to
the output stage.
In practice, for hi-fi use the amp has no bad habits due
to the cap coupling. The gate input impedance of the mosfets is extremely high
like the grids of a tube, and hence CR couping is a very effective coupling
method, and just because solid state devices are used there is no reason why
direct coupling must be used as in the 2 x 300w amp above.
The low frequency
stability is not compromised since there are so few consecutive stages. The use
of the OPT does not permit direct coupling of the mosfets to the output, so the
failure of a mosfet will prevent any connection of the internal DC supply to any
speaker connected.
No protection circuit is needed apart from
fuses.
Each mosfet under the above operating condition acts with a similar
distortion profile to a good beam tetrode, but at low power, they are
surprisingly linear. Since mosfets do not use current to drive them and are
voltage operated devices, they can be thought of like tubes.
In this application, I measured the mosfets to have
amplification factor, µ = 180,
Dynamic drain resistance, Rd = 220 Ohms,
and
Transconductance, gm = 0.8 amps per volt.
The calculation of gain is the same as for a tube.
Voltage gain
= µ x RL = 180 x
17.5 = 24.7 in
this case.
RL + Rd 17.5 +
220
The mosfets are set up like pentode tubes in the circuit, and have
the load connected in the drain circuit, and the effective drain to drain source
resistance with the 0.22 ohms measures about 260 ohms, for the four mosfets
together. Since the OPT has an impedance ratio of 35 to 5 ohms, or 7 : 1, the
Rd-d is transformed down to 260/7 = 37 ohms at the speaker output terminals,
which is far to high to drive speakers effectively.
It is very similar to the
case where 2 x EL34 are set up in class A pentode with an OPT. Ra-a is higher,
but so is the OPT ZR, and the Rout at the secondary is still many times the load
resistance and the damping factor which = RL / Rsource
is too low to keep
the speaker and its filters free of gross frequency distortion, ie, with high
Rout greater than RL
one gets gross variations in acoustic levels at various
frequencies. All loudspeakers are designed for amplifiers with Rout
of less
than 1 ohm.
Since the pushpull class A action has only a small amount of distortion at a few watts, the only reason to use FB is to reduce output impedance and any noise in the circuit. In these regards the solid state is no better than tubes, and in fact not as good as triodes which require no added external loops of NFB if the designer is careful.
There are 3 feedback loops in the PP50 mosfet amps.
The first two are Balanced Shunt
Feedback loops from the primary of the OPT ( the mosfet drain connections
) back to the collector circuit of the LTP driver transistors via the 22k shunt
feedback resistors, R16/R19. Since the voltage at the gate is -0.9v for +20v at
the drain, the current in the 22 k R is nearly 1mA, and the load the driver
transistors effectively see is only 900 ohms,
because RL = V / I = 0.9 /
0.001 = 900 ohms.
The driver stage has a gain determined approximately by its gm x RL.
The driver stage transistors form a balanced
signal current source because
of the high collector resistance.
If there is a reduction in mosfet gain say due to a lower load being
used, the effective load seen by the driver stage increases. For example, if the
load is halved from 5 ohms to 2.5 ohms at the output, then 0.9Vrms applied
to the mosfet gates would produce only 10Vrms at each drain instead of 20Vrms.
The current flow through the shunt FB resistors
would reduce to nearly half,
so the load seen by driver stage would nearly double, and as the gain of the
driver stage is about proportional to load, a larger voltage would try to appear
at the driver collectors immediately with a lower load
value, and hence the
mosfet drain to drain output voltage would tend to be maintained regardless of
load values
at the OPT secondary.
So any drop in gain of the mosfets due say to a drop in load value
will be compensated for, and in fact with just these two balanced loops of shunt
NFB, the Rout is reduced from 37 Ohms to about 1 ohm at the output.
Any
distortion voltage appearing at the mosfet drain will be divided by the 22k and
4.7k collector dc supply resistors,
R16 / R10.
ß, the fraction fed back =
4.7 / ( 4.7 + 22 ).
If the thd at a drain was +0.1Vrms, then +0.0176Vrms is
being applied at the gate. This is amplified
about 20 times by the mosfet to
produce -0.352Vrms. This subtracts from the distortion voltage of +0.452Vrms
that must have been present without NFB applied.
The amount of locally
applied NFB is about 13 dB.
In actual fact the dual shunt FB loops applied around each pair of
output mosfets acts similarly to the electrostatic
shunt feedback acting
inside the triode, except I am using resistances to apply the local feedback.
I doubt that anyone understood a single word of what I just tried to
say simply in a few paragraphs about shunt NFB.
So let me see if you get the
message any clearer when i try to confuse you further with a diagram of a
basic
shunt NFB loop with single 2SK134 driven by a an ordinary small signal
npn bjt.
Unfortunately, I cannot explain things like amplifiers any more
simply because they simply contain a lot
things all interacting
simultaneously, and to know, you must look, because if you don't look yer won't
know!
Fig 
In the schematic, I have drawn up the bjt and mosfet as equivalent
generator models with a triangle to represent
an internal low output
resistance voltage generator with a phase inverting output of gm x Rc or
gm x Rd.
The bjt can be drawn as a model gene producing input voltage x
gm x Rc, ( Rc is collector resistance ).
In this case a common little bjt may
give gm x Rc = 1,000. The product of gm x Rc is actually like the product
of
gm Ra for tubes, and is = amplification factor, µ, but we don't speak of bjts
having a µ. Such is engineering conventions.
The presence of an emitter
resistor, Re, of 50 ohms will have considerable effect on the Rout at the
point of the triangle
which is the gene output. The gene does not actually
exist, nor do te voltages at the output of the gene, but the gene is a good equivalent model to use for basic
devices!
The collector resistance acts like a series R between gene
and the load, and the actual collector terminal C
is shown to the right of
Rc, 50k.
At this point C in this case, the effective output resistance of the
bjt set up as shown is 100k.
Now the model for the 2SK134 mosfet is similar, with a triangle gene
producing an output of Vg-s x gm x Rd.
The 0.22 source resistor is shown. The
load of 35 ohms has + 20Vrms signal.
Instaneous phases of the relative signal phases are shown with + or
- signs. To analyse the operation of the circuit
and draw all the current
flows around it, one can strat with the load current and work backwards towards
the input.
This analysis is the most basic type of current flow lesson that
should be known by anyone building amplifiers.
The input signal produced by the input bjt circuit is from a 100k
sourse resistance.
The collector of the bjt has a dc supply R plus biasing R for the
gate of the mosfet and thse parallel R are shown as one R,
R1FB = 4k4. The
R2FB is the resistor from the output back to the gate and collector
junction.
From looking at the voltages, you can see that the circuit
resembles a single ended version of the PP 50 watt mosfet
class A amplifier,
and it is intended this way, to allow fuller understanding, and perhaps lead you
to designing your own
simple SE amp using a mosfet and a bjt
driver.
Looking at the network of R around the mosfet, let us suppose a
distortion voltage +0.2V appears at the load due to the mosfet's tendency to add
harmonics not in the input signal.
This signal produces a current flow back
towards the input of the amp via R2FB and then R1FB in parallel with the
effective Rout of the bjt output resistance of 100k. So we have a total R =
22k + 100k//4k4 = 26k2
From Ohm's law we can say that distortion current flow
= 0.2 / 26.2 = 0.00763mA.
It doesn't sound like much current, but it is
there.
So the Vd at the gate, G, in our circuit = Id x 100k//R1FB = 0.00763 x
4.2k = +0.032V.
This voltage is amplified by the mosfet gain including the
effect of the source R which is 20V / 0.935 = 21.4.
The phase of the Vd is inverted, and an output drain voltage appears
as - 21.9 x 0.032V = - 0.7V.
How the hell can this be? we just said that we
measured and began with +0.2V of distortion at the load!!!!
Well, the mosfet
without NFB was already making a considerably larger +Vd, but when an additional
-Vd is
introduced by the use of NFB, the +Vd without NFB is reduced to only
+0.2V.
So commonsense tells us that the Vd must have been 0.9V without any
NFB.
Add feedback, and viola, Vd reduces from +0.9V to
0.2V.
This reduction amounts to about 13dB of
NFB.
The term Shunt voltage NFB is because the inverted output voltage
signal is directly fed back by resiostor dividers to oppose the input signal of
an amp. Series voltage NFB is the more common NFB and the resistor
dividers
apply a fed back signal to a point in the input circuit which has
the same phase as the input, and so the FB is
"in series" with the input
sugnal which always must be a higher amplitude than the FB
signal.
In the case of the emitter resistor in the bjt circuit and source R
in the mosfet circuit there is Series current NFB
where there is a voltage
generated by the output load current flowing in a resistor and which appears in
series to the
input base or gate signal.
If we return to the workings of the PP50 mosfet amp, the global FB
network
is a sample of series voltage NFB. Those of you with real
understanding so far could see that a loop
of global series voltage NFB could
be added by connection of a resistance lbetween the 35ohm output lod signal
and the emitter resistance of the bjt input
device.
The bjt in the modelling shows that the current in the bjt and its
50k of Rc is equal to the total of currents in
R1FB and R2FB, and = 1.167
mA.
Since the voltage change at G = 0.935V, the load seen by the collector at
point C = 0.935 / 1.167 = 0.8k ohms.
A small signal input transistor like
this has in effect a load of 800 ohms with an emitter R = 50 and its gain
is
0.935 / 0.176V = 5.31, reduced from 0.935 / 0.059 = 15.8 without the Re.
In
fact if we made Re much larger we could have the input bjt operating with about
1V input, and the local NFB in the bjt
would be higher and so would its thd,
a cause for some concern, because a single bjt has maybe 20 times the thd of a
triode used in a similar function. But the global NFB could be used to apply
about 0.6V NFB at the bjt emitter
which would reduce the thd to good enough
levels.
Of course just one 2SK134 set up with supply = 34V and Id = 0.8 amps
is only good for 11 watts into 35ohms.
The operating point of 34V x 0.8A
gives 27.2 watts of idle dissipation. This is a safe figure to begin with. Nelso
Pass
in his Zen amps has a single mosfet with 17V x 3A at idle for = 51 watts
dissipation and a freind of mine
tried to build a Zen and after ruining 3
expensive IRF mosfets due to inexplicable "smoke and silence" events he gave
up
and has never found the time to learn more and complete the amp. Take my
word for it, 30 watts is the absolute
limit for idle dissipation in any
class A flat pack or TO3 devices.
The operating condition suits a 35 ohm load
and gives the mosfet plenty of gain and moderate thd compared
to where one
may have lower V and higher I to suit a lower load
value.
But to get 44 watts into
8.75 ohms, 4 mosfets in parallel are needed, or 6 for 5.8
ohms.
If one runs the mosfet with 20V supply, then Ia can be 1.25A for 25
watts dissipation and you can get
about 9.4W into 12ohms, or with two
mosfets, 18.8W into 6 ohms and so on,
but mosfet gain is much less at only
9.6 with a 12 ohm load so the shunt NFB is less
effective.
In the PP50 I found the noise was still a little high, and so I applied 12 dB of global loop feedback from the secondary of the OPT back to the second available input of the LTP, and the Rout fell to a satisfactory 0.2 Ohms, and the noise vanished, and the THD was less than 0.2% at a dB below clipping, and declining towards zero as the output power was reduced. There is absolutely ZERO crossover distortion.
The amp needed zobel networks across the collectors of the LTP, and across each half of the OPT, to shelve the HF gain and control phase shift, in exactly the same manner as a tube amp would. The OPT has a full power bandwidth from 10Hz to 300kHz, ( yes, not a typo, it is 300kHz ) but this much bandwidth cannot be used, and to control overshoot on square waves with capacitor loads above 40 kHz, the shelving networks were employed, and so the final finished bandwidth is a more sensible 10Hz to 65kHz.
About 99.9% of mainstream solid state amps will not use an OPT. I
have proved to myself at least that they can be used effectively and my mosfet
amp has the ability to get a good load match even to 4 ohms, and still get class
A sound, and loads of it.
Since all the output devices are NPN types with
the same part number, the even order distortion of the devices on each side of
the PP circuit cancel almost perfectly.
The circuit doesn't need an
intermediary high gain voltage amplification stage.
Instead of having a
single phase of drive signal applied to a complementary PP pair of PNP and NPN
devices,
the two output phases of the input differential amp are both
used.
The difference between PNP and NPN devices when in common source mode
is substantial, equivalent to having
an EL34 on one side of a tube PP amp and
a 6L6GC on the other; it does not give the best
outcome.
Improvements to the amplifier may result by using a pair of matched
darlington pairs for the input/driver LTP, which would increase their local open
loop gain and linearity considerably, and increase the input impedance of the
amp slightly.
But after having used 2SK369 j-fets in phono stages I have wanted to
use 3 of these for the input LTP and CCS
to make the amp a totally fet
amplifier.
The 2SK369 is exactly the same as 2SK147 ( which is no longer made
by Hitachi ) except slightly lower Pd rating. It has high input impedance and gm
= 40mA / V at 5mA of idle current and would suit this application perfectly.
They are very low noise devices, much quieter than any bjt with the same idle
current.
I'm still looking for the time to change over to full fet
topology....
Most tube amps are terribly simple in their circuits, and the simplest will use just four active elements, the two halves of the output stage, and a pair of input tubes, to give the driver stage some gain, and provide a balanced drive to the output tubes. Quad II is a sample of what I mean.
Most modern 50 watt SS amps will have up to 25 active elements all doing their thing simultaneously ( in theory ) to produce the sound. I wanted to keep to using 4 elements. The CCS tail of the LTP could be considered a fifth element, but its action is really part of the power supply, and its function is to provide a constant current to the diff amp emitter circuit. The circuit has very good CMRR, so there is no need for any better regulation of the supply voltages than the simple zeners shown.
The drain supply part of the power supply uses two 4,700 uF input caps, where about 0.7 vrms of ripple voltage is produced by 3 amps of DC draw. Then there is a low DC resistance choke of 15 mH using thick wire, which then feeds 3 x 15,000 uF caps. The ripple voltage at the CT is thus reduced 269 times to only 2.6 mV, which is low enough so that if loads below 4 Ohms are used, then the crossover to class AB will be unmolested by power supply ripple entering the amp.
The voltage supplies for the driver stage are higher than the main drain supply, +/- 65 volts, and this allows for regulation down to suitable lower voltages at low current for the class A driver amp to work from. The amp is not troubled by wide variations in the mains voltages.
The heatsink for the amp was DIY from various pieces of heavy angle
section of aluminium I had picked up over the years. I needed something which
could easily dissipate 100 watts continuously. I found that placing all four
mosfets close together on an angle which then connected to the main body of the
sink means that the centre part of the sink gets warmer than the ends but
nevertheless the devices stay cool enough. The heat sink I used is 350 mm long,
120 high, with 20 fins 70 mm x 120 x 1.6 mm thick. The fins and bars used in the
heatsink were just bolted together with plenty of white heat conductive
paste between mating surfaces.
The secret for a good heatsink is to have plenty of fin area, have vertical fins to allow air to flow easily
upwards as it
warms and to have the sink exposed to an unfttered air flow.
The Quad 405 used this idea well with an external heatsink that is large and
rugged. Anyone building such an amp could use a ready made extruded
heatsink. There are standard
extruded heatsinks available which are
300mm long, 150mm high, with 30 fins 40mm long by 2mm thick
which are good
for about 100 watts dissipation without a fan but to be sure the 200mm high
sink
would be the best. Sugden class A SS amps show how not to do it properly by having their
heatsinks
with the fins flat and running horizontal and with poor access to
devices. Boy they get hot during
Australia's summer weather!
The
extruded heatsink with one flat side allows for simple bolting of flatpack
mosfets more easily than all the fiddling I went to mount a quad of TO3 on an
angle shelf in the middle of the amp. The angle is 6mm thickness so there was
much drilling.
In the other mosfet based stereo amp with 350 watts per channel
capability, each channel draws 40 watts at idle, which barely warms the amp. For
each channel, it uses a the extruded heat sink 300 mm long, 150 mm high, with 30
fins 40 mm x 150 mm x 2 mm thick, and I placed the 6 mosfets right along the
sink, and thus the temperature is lower and evenly spread right along the sink.
There is room for two more mosfets at each end of the sink if need
be.
During the years I have used these amps, nobody has complained about the sound, which nobody can distinguish from tube amps in AB testing, where the speakers are switched from a class A tube amp to the mosfet amp.
Over longer
testing periods, I would say the tubes still have a slight
lead.
There are
other ways to use mosfets for class A SE amps.
One way is to have 4 parallel
NPN mosfets with a 0.16H inductor between the drain and a +22V supply and
with idle current = 4.5 amps for total device dissipation of 100 watts. The
inductor must have its wire size large enough so the current density is less
than 2A/sq.mm to give a cool running coil. See my pages on power supply and
choke design.
The load can be coupled to the drain using four paralleled
4,700 uF high ripple current rated electro caps
rated for 50V at least.
This will not sound any worse than using an SE
OPT .
Sugden have been using capacitor coupling to output loads for
40 years with rave reviews,
and you can't roast a speaker voice coil with dc
from the power supply. Such a simple output stage
will produce a maximum of
about 40 watts of pure class A into 4 ohms, or about 23 watts into 8
ohms.
Should one
want to use no inductors and settle for a complementary pair PP output stage and
driver stage very
similar to the 300 watt AB stereo amp above, then with 3
npn and 3 pnp mosfets and supply rails at +/-22V
and idle current = 2.27Adc,
then you would have only 16.6W of dissipation per device, and still be able
to
to get a +/- 4.5amp peak current swing into 4 ohms for 40 watts of class
A.
The power into all loads above 4 ohms will be pure class A but limited by
the available voltage swing of about
+/- 20 peak volts maximum.
But where
loads are lower than 4 ohms the power will be class AB and into 2 ohms the max
AB power
will be 80 watts and class A power will still be 20 watts.
Since the driver stage has only to produce a maximum of about 25vrms to
drive the output stage
the 300 watt amp schematic could be used with say
+/-50 rails fro the driver amplifier.
It would give blamless performance with
very low thd and excellent sound.
As I discussed above, the main amount
of distortion is produced in the driver stages. The class A
complementary
source follower pair will have less than 0.4% thd at full power into 6 ohms, and
zero crossover distortion and Rout < 1 ohm, so without any global NFB the
output stage
is as good as many tube amps with 20dB of global NFB. The NFB
is in the source follower connection.
One could build a bjt driver amp
as I have above with the 2 x 300w with low thd by having a loop of FB from
the MJE340/350 collectors back to the differential input amp NFB input. The
set up of the MJE340/350
gain pair would require that the NFB resistance
network load the bjts to reduce their open loop gain somewhat
so that a lower
amount of NFB is actually used.
I am sure the
brighter mortals amoung you will tailor your diy efforts to suit yourselves
without smoke and
give your ears good
music.
There are
1,001 ways to build an amplifier.
This shows
the inside of the 50 watt monos. The psu torodial power transformer
and choke
are towards the left behind two 4,700uF caps, circuit board in the
centre,
more psu caps and 5Kg OPT with C cores at the rear
right.