The input stage consists of a couple of general purpose
small signal transistors which are arranged as a longtail pair amp,
(differential pair amp), with a low noise BC109 as a constant current
source for the common emitter circuit. The two halves of the LTP have
some local current FB via the 47 ohm emitter resistors, R11,12. The
LTP is quite linear in its operation, and only needs to generate up to
about 1Vrms of signal at each collector to produce full power at the
output.
The output stage uses four N type Hitachi output mosfets
which are fed with a +35 volt 3 amp supply. There is a wide bandwidth
output transformer using C-cores, about the same size as a 50 watt OPT
on a tube amp to couple the mosfets to the load. The OPT has a ratio of
35:5 ohms, and has thicker wire than a tube amp in the primary, but has
a 5P x 6S interleaving patteren to get a 300kHz HF cut off. The dual
C-cores have a small air gap to reduce the effective µ to about
2,200 to prevent saturation effects below 16Hz at full power.
Since mosfets can generate much higher peak current than any tube, the
option of air gapping the PP core
resulted in much reduced current spikes below 16Hz at full power. This
opened my mind to using gapped cores
cores for any PP transformer. With modern GOSS core material the
µ of the core may be much higher than it needs to be
and a reduction with an air gap may not substantially reduce the
primary inductance that shunts the RL and Ra-a
or the RL and Rd-d, the drain to drain resistance of the mosfets.
Modern GOSS I use with E&I lams maximally
intermeshed will give a µ max = 17,000, and it could be 1,700
without causing any loss of response
at bass frequencies above 14 Hz.
Each mosfet has an idle current of about 0.7 amps, and
dissipates 24 watts each, (like an EL34).
This is the reliable continuous limit for a TO3 package device imho.
The drain to
drain load is 35 ohms, so each half of the PP circuit has a 17.5 ohm
load in class A, and thus each mosfet sees 35 ohms of load. Each
mosfet is biased with 0.8 amps
of idle current, and there is a peak current swing in each mosfet of
+/- 0.8 amps peak. The voltage swing at the drain is about +/- 20V
peak, so each mosfet produces 11.2 watts, which makes a total of 44.8
watts class A for the 4 mosfets.
The mosfets is biased in their negative temperature
coeficient region which means that if they heat up the bias current
tends to reduce. Unlike power transistors, they don't run away
thermally due to the positive temp coefficient.
Therefore the mosfets may have have truly
fixed bias set by the resistive divider of R21/R22, 32k and 1k,
providing about +1volt bias to the gates via the 68 k bias
resistors. There is cap coupling from the driver stage, but because
there are only two stages in the is amp, the stability at LF is
excellent. At gross overload, the caps charge up like they do on a tube
amp,
and the mosfets are driven into a temporary state of harmless cut off.
This only occurs when the power exceeds the
clipping level, and there is zero need to worry about the effect under
normal operation, just as is the case with a good tube amp.
The 7.5 volt back-to-back zeners control the applied voltages
to the output stage.
In practice, for hi-fi use the amp has no bad habits due to the cap
coupling. The gate input impedance of the mosfets is extremely high
like the grids of a tube, and hence CR couping is a very effective
coupling method, and just because solid state devices are used there is
no reason why direct coupling must be used as in the 2 x 300w amp above.
The low
frequency stability is not compromised since there are so few
consecutive stages. The use of the OPT does not
permit direct coupling of the mosfets to the output, so the failure of
a mosfet will prevent any connection of the internal DC supply to any
speaker
connected.
No protection circuit is needed apart from fuses.
Each mosfet under the above operating condition acts
with a similar distortion profile to a good beam tetrode, but at low
power, they are surprisingly linear. Since mosfets do not use current
to drive them and are voltage operated devices, they can be thought of
like tubes.
In this application, I measured the
mosfets to have
amplification factor, µ = 180,
Dynamic drain resistance, Rd = 220 Ohms, and
Transconductance, gm = 0.8 amps per volt.
The calculation of gain is the same as for a tube.
Voltage
gain =
µ x
RL
= 180 x
17.5 =
24.7 in this case.
RL + Rd 17.5 + 220
The mosfets are set up like pentode tubes in the
circuit, and have the load connected in the drain circuit, and the
effective drain to drain source resistance with the 0.22 ohms measures
about 260 ohms,
for the four mosfets together. Since the OPT has an impedance ratio of
35 to 5 ohms, or 7 : 1, the Rd-d is transformed down to 260/7 = 37 ohms
at the
speaker output terminals, which is far to high to drive speakers
effectively.
It is very similar to the case where 2 x EL34 are set up in class A
pentode with an OPT. Ra-a is higher, but so is the OPT ZR, and the Rout
at the secondary is still many times the load resistance and the
damping factor which = RL / Rsource
is too low to keep the speaker and its filters free of gross frequency
distortion, ie, with high Rout greater than RL
one gets gross variations in acoustic levels at various frequencies.
All loudspeakers are designed for amplifiers with Rout
of less than 1 ohm.
Since the pushpull class A action has only a small amount of distortion at a few watts, the only reason to use FB is to reduce output impedance and any noise in the circuit. In these regards the solid state is no better than tubes, and in fact not as good as triodes which require no added external loops of NFB if the designer is careful.
There are 3 feedback loops in the PP50 mosfet amps.
The first two are Balanced
Shunt Feedback loops from the primary
of the OPT ( the mosfet drain connections ) back to the collector
circuit of the LTP driver transistors via the 22k shunt feedback
resistors, R16/R19. Since the voltage at the gate is -0.9v for +20v at
the drain, the current in the 22 k R is nearly 1mA, and the load the
driver transistors effectively see is only 900 ohms,
because RL = V / I = 0.9 / 0.001 = 900 ohms.
The driver stage has a gain determined approximately by
its gm x RL. The driver stage transistors form a balanced
signal current source because of the high collector resistance.
If there is a reduction in mosfet gain say due to a
lower load being used, the effective load seen by the driver stage
increases. For example, if the load is halved from 5 ohms to 2.5
ohms at the output, then 0.9Vrms applied to the mosfet gates would
produce only 10Vrms at each drain instead of 20Vrms. The current flow
through the shunt FB resistors
would reduce to nearly half, so the load seen by driver stage would
nearly double, and as the gain of the driver stage is about
proportional to load, a larger voltage would try to appear at the
driver collectors immediately with a lower load
value, and hence the mosfet drain to drain output voltage would tend to
be maintained regardless of load values
at the OPT secondary.
So any drop in gain of the mosfets due say to a drop in
load value will be compensated for, and in fact with just these two
balanced loops of shunt NFB, the Rout is reduced from 37 Ohms to about
1 ohm at the output.
Any distortion voltage appearing at the mosfet drain will be divided by
the 22k and 4.7k collector dc supply resistors,
R16 / R10.
ß, the fraction fed back = 4.7 / ( 4.7 + 22 ).
If the thd at a drain was +0.1Vrms, then +0.0176Vrms is being applied
at the gate. This is amplified
about 20 times by the mosfet to produce -0.352Vrms. This subtracts from
the distortion voltage of +0.452Vrms
that must have been present without NFB applied.
The amount of locally applied NFB is about 13 dB.
In actual fact the dual shunt FB loops applied around
each pair of output mosfets acts similarly to the electrostatic
shunt feedback acting inside the triode, except I am using resistances
to apply the local feedback.
I doubt that anyone understood a single word of what I
just tried to say simply in a few paragraphs about shunt NFB.
So let me see if you get the message any clearer when i try to confuse
you further with a diagram of a basic
shunt NFB loop with single 2SK134 driven by a an ordinary small signal
npn bjt.
Unfortunately, I cannot explain things like amplifiers any more simply
because they simply contain a lot
things all interacting simultaneously, and to know, you must look,
because if you don't look yer won't know!
Fig
In the schematic, I have drawn up the bjt and mosfet as
equivalent generator models with a triangle to represent
an internal low output resistance voltage generator with a phase
inverting output of gm x Rc or gm x Rd.
The bjt can be drawn as a model gene producing
input voltage x gm x Rc, ( Rc is collector resistance ).
In this case a common little bjt may give gm x Rc = 1,000. The product
of gm x Rc is actually like the product
of gm Ra for tubes, and is = amplification factor, µ, but we
don't speak of bjts having a µ. Such is engineering conventions.
The presence of an emitter resistor, Re, of 50 ohms will have
considerable effect on the Rout at the point of the triangle
which is the gene output. The gene does not actually exist, nor do te
voltages at the output of the gene, but the gene is a good equivalent model to use for basic
devices!
The collector resistance acts like a series R between gene and the
load, and the actual collector terminal C
is shown to the right of Rc, 50k.
At this point C in this case, the effective output
resistance of the bjt set up as shown is 100k.
Now the model for the 2SK134 mosfet is similar, with a
triangle gene producing an output of Vg-s x gm x Rd.
The 0.22 source resistor is shown. The load of 35 ohms has + 20Vrms
signal.
Instaneous phases of the relative signal phases are
shown with + or - signs. To analyse the operation of the circuit
and draw all the current flows around it, one can strat with the load
current and work backwards towards the input.
This analysis is the most basic type of current flow lesson that should
be known by anyone building amplifiers.
The input signal produced by the input bjt circuit is
from a 100k sourse resistance.
The collector of the bjt has a dc supply R plus biasing
R for the gate of the mosfet and thse parallel R are shown as one R,
R1FB = 4k4. The R2FB is the resistor from the output back to the gate
and collector junction.
From looking at the voltages, you can see that the circuit resembles a
single ended version of the PP 50 watt mosfet
class A amplifier, and it is intended this way, to allow fuller
understanding, and perhaps lead you to designing your own
simple SE amp using a mosfet and a bjt driver.
Looking at the network of R around the mosfet, let us
suppose a distortion voltage +0.2V appears at the load due to the
mosfet's tendency to add harmonics not in the input signal.
This signal produces a current flow back towards the input of the amp
via R2FB and then R1FB in parallel with the
effective Rout of the bjt output resistance of 100k. So we have a total
R = 22k + 100k//4k4 = 26k2
From Ohm's law we can say that distortion current flow = 0.2 / 26.2 =
0.00763mA.
It doesn't sound like much current, but it is there.
So the Vd at the gate, G, in our circuit = Id x 100k//R1FB = 0.00763 x
4.2k = +0.032V.
This voltage is amplified by the mosfet gain including the effect of
the source R which is 20V / 0.935 = 21.4.
The phase of the Vd is inverted, and an output drain
voltage appears as - 21.9 x 0.032V = - 0.7V.
How the hell can this be? we just said that we measured and began with
+0.2V of distortion at the load!!!!
Well, the mosfet without NFB was already making a considerably larger
+Vd, but when an additional -Vd is
introduced by the use of NFB, the +Vd without NFB is reduced to only
+0.2V.
So commonsense tells us that the Vd must have been 0.9V without any NFB.
Add feedback, and viola, Vd reduces from +0.9V to 0.2V.
This reduction amounts to about 13dB of NFB.
The term Shunt voltage NFB is because the inverted
output voltage signal is directly fed back by resiostor dividers to
oppose the input signal of an amp. Series voltage NFB is the more
common NFB and the resistor dividers
apply a fed back signal to a point in the input circuit which has the
same phase as the input, and so the FB is
"in series" with the input sugnal which always must be a higher
amplitude than the FB signal.
In the case of the emitter resistor in the bjt circuit
and source R in the mosfet circuit there is Series current NFB
where there is a voltage generated by the output load current flowing
in a resistor and which appears in series to the
input base or gate signal.
If we return to the workings of the PP50 mosfet amp, the
global FB network
is a sample of series voltage NFB. Those of you with real
understanding so far could see that a loop
of global series voltage NFB could be added by connection of a
resistance lbetween the 35ohm output lod signal
and the emitter resistance of the bjt input device.
The bjt in the modelling shows that the current in the
bjt and its 50k of Rc is equal to the total of currents in
R1FB and R2FB, and = 1.167 mA.
Since the voltage change at G = 0.935V, the load seen by the collector
at point C = 0.935 / 1.167 = 0.8k ohms.
A small signal input transistor like this has in effect a load of 800
ohms with an emitter R = 50 and its gain
is 0.935 / 0.176V = 5.31, reduced from 0.935 / 0.059 = 15.8 without the
Re.
In fact if we made Re much larger we could have the input bjt operating
with about 1V input, and the local NFB in the bjt
would be higher and so would its thd, a cause for some concern, because
a single bjt has maybe 20 times the thd of a triode used in a similar
function. But the global NFB could be used to apply about 0.6V NFB at
the bjt emitter
which would reduce the thd to good enough levels.
Of course just one 2SK134 set up with supply = 34V and
Id = 0.8 amps is only good for 11 watts into 35ohms.
The operating point of 34V x 0.8A gives 27.2 watts of idle dissipation.
This is a safe figure to begin with. Nelso Pass
in his Zen amps has a single mosfet with 17V x 3A at idle for = 51
watts dissipation and a freind of mine
tried to build a Zen and after ruining 3 expensive IRF mosfets due to
inexplicable "smoke and silence" events he gave up
and has never found the time to learn more and complete the amp. Take
my word for it, 30 watts is the absolute
limit for idle dissipation in any class A flat pack or TO3 devices.
The operating condition suits a 35 ohm load and gives the mosfet plenty
of gain and moderate thd compared
to where one may have lower V and higher I to suit a lower load value.
But to get 44 watts into
8.75 ohms, 4 mosfets in parallel
are needed, or 6 for 5.8 ohms.
If one runs the mosfet with 20V supply, then Ia can be
1.25A for 25 watts dissipation and you can get
about 9.4W into 12ohms, or with two mosfets, 18.8W into 6 ohms and so
on,
but mosfet gain is much less at only 9.6 with a 12 ohm load so the
shunt NFB is less effective.
In the PP50 I found the noise was still a little high, and so I applied 12 dB of global loop feedback from the secondary of the OPT back to the second available input of the LTP, and the Rout fell to a satisfactory 0.2 Ohms, and the noise vanished, and the THD was less than 0.2% at a dB below clipping, and declining towards zero as the output power was reduced. There is absolutely ZERO crossover distortion.
The amp needed zobel networks across the collectors of the LTP, and across each half of the OPT, to shelve the HF gain and control phase shift, in exactly the same manner as a tube amp would. The OPT has a full power bandwidth from 10Hz to 300kHz, ( yes, not a typo, it is 300kHz ) but this much bandwidth cannot be used, and to control overshoot on square waves with capacitor loads above 40 kHz, the shelving networks were employed, and so the final finished bandwidth is a more sensible 10Hz to 65kHz.
About 99.9% of mainstream solid state amps will not use
an OPT. I have proved to myself at least that they can be used
effectively and my
mosfet amp has the ability to get a good load match even to 4 ohms, and
still get class A sound, and loads of it.
Since all the output devices are NPN types with the same part number,
the even order distortion of the devices on each side of the PP circuit
cancel almost perfectly.
The circuit doesn't need an intermediary high gain voltage
amplification stage.
Instead of having a single phase of drive signal applied to a
complementary PP pair of PNP and NPN devices,
the two output phases of the input differential amp are both used.
The difference between PNP and NPN devices when in common source mode
is substantial, equivalent to having
an EL34 on one side of a tube PP amp and a 6L6GC on the other; it does
not give the best outcome.
Improvements to the amplifier may result by using a pair
of matched darlington pairs for the input/driver LTP, which would
increase their local open loop gain and linearity considerably, and
increase the input impedance of the amp slightly.
But after having used 2SK369 j-fets in phono stages I
have wanted to use 3 of these for the input LTP and CCS
to make the amp a totally fet amplifier.
The 2SK369 is exactly the same as 2SK147 ( which is no longer made by
Hitachi ) except slightly lower Pd
rating. It has high input impedance and gm = 40mA / V at 5mA of idle
current and would suit this application
perfectly. They are very low noise devices, much quieter than any bjt
with the same idle current.
I'm still looking for the time to change over to full fet topology....
Most tube amps are terribly simple in their circuits, and the simplest will use just four active elements, the two halves of the output stage, and a pair of input tubes, to give the driver stage some gain, and provide a balanced drive to the output tubes. Quad II is a sample of what I mean.
Most modern 50 watt SS amps will have up to 25 active elements all doing their thing simultaneously ( in theory ) to produce the sound. I wanted to keep to using 4 elements. The CCS tail of the LTP could be considered a fifth element, but its action is really part of the power supply, and its function is to provide a constant current to the diff amp emitter circuit. The circuit has very good CMRR, so there is no need for any better regulation of the supply voltages than the simple zeners shown.
The drain supply part of the power supply uses two 4,700 uF input caps, where about 0.7 vrms of ripple voltage is produced by 3 amps of DC draw. Then there is a low DC resistance choke of 15 mH using thick wire, which then feeds 3 x 15,000 uF caps. The ripple voltage at the CT is thus reduced 269 times to only 2.6 mV, which is low enough so that if loads below 4 Ohms are used, then the crossover to class AB will be unmolested by power supply ripple entering the amp.
The voltage supplies for the driver stage are higher than the main drain supply, +/- 65 volts, and this allows for regulation down to suitable lower voltages at low current for the class A driver amp to work from. The amp is not troubled by wide variations in the mains voltages.
The heatsink for the amp was DIY from various pieces of
heavy angle section of aluminium I had picked up over the years. I
needed something which could easily dissipate 100 watts continuously. I
found that placing all four mosfets close together on an angle which
then connected to the main body of the sink means that the centre part
of the sink gets warmer than the ends but nevertheless the devices stay
cool enough. The heat sink I used is 350 mm
long, 120 high, with 20 fins 70 mm x 120 x 1.6 mm thick. The fins and
bars used in the heatsink were just bolted together with plenty of
white heat conductive
paste between mating surfaces.
The secret for a good heatsink is to have plenty of fin
area, have vertical fins to
allow air to flow easily upwards as it
warms and to have the sink exposed to an unfttered air flow. The Quad
405 used this idea well with an external heatsink that is large and
rugged. Anyone building such an amp could use a ready made extruded
heatsink. There are standard
extruded heatsinks available which are 300mm long, 150mm high, with 30
fins 40mm long by 2mm thick
which are good for about 100 watts dissipation without a fan but to be
sure the 200mm high sink
would be the best. Sugden class A SS amps show how not to do it properly by
having their heatsinks
with the fins flat and running horizontal and with poor access to
devices. Boy they get hot
during Australia's summer weather!
The extruded heatsink with one flat side allows for simple bolting of
flatpack mosfets
more easily than all the fiddling I went to mount a quad of TO3 on
an angle shelf in the middle of the amp. The angle is 6mm thickness so
there was much drilling.
In the other mosfet based stereo amp with 350 watts per
channel capability, each channel draws 40 watts at idle, which barely
warms
the amp. For each channel, it uses a the extruded heat sink 300 mm
long, 150 mm high, with 30 fins 40 mm x 150 mm x 2 mm thick, and I
placed the 6 mosfets right along the sink, and thus the temperature is
lower and evenly spread right along the sink. There is room for two
more mosfets at each end of the sink if need be.
During the years I have used these amps, nobody has complained about the sound, which nobody can distinguish from tube amps in AB testing, where the speakers are switched from a class A tube amp to the mosfet amp.
Over longer testing
periods, I would say the tubes still have a slight lead.
There are other ways
to use mosfets for class A SE amps.
One way is to have 4 parallel NPN mosfets with a 0.16H inductor
between the drain and a +22V supply and with idle current = 4.5 amps
for total device dissipation of 100 watts. The inductor must have its
wire size large enough so the current density is less than 2A/sq.mm to
give a cool running coil. See my pages on power supply and choke design.
The load can be coupled to the drain using four paralleled 4,700 uF
high ripple current rated electro caps
rated for 50V at least. This will
not sound any worse than using an SE OPT .
Sugden have been using capacitor coupling to output loads for 40 years
with rave reviews,
and you can't roast a speaker voice coil with dc from the power supply.
Such a simple output stage
will produce a maximum of about 40 watts of pure class A into 4 ohms,
or about 23 watts into 8 ohms.
Should one want to
use no inductors and settle for a complementary pair PP output stage
and driver stage very
similar to the 300 watt AB stereo amp above, then with 3 npn and 3 pnp
mosfets and supply rails at +/-22V
and idle current = 2.27Adc, then you would have only 16.6W of
dissipation per device, and still be able to
to get a +/- 4.5amp peak current swing into 4 ohms for 40 watts of
class A.
The power into all loads above 4 ohms will be pure class A but limited
by the available voltage swing of about
+/- 20 peak volts maximum.
But where loads are lower than 4 ohms the power will be class AB and
into 2 ohms the max AB power
will be 80 watts and class A power will still be 20 watts.
Since the driver stage has only to produce a maximum of about 25vrms to
drive the output stage
the 300 watt amp schematic could be used with say +/-50 rails fro the
driver amplifier.
It would give blamless performance with very low thd and excellent
sound.
As I discussed above, the main amount of distortion is produced
in the driver stages. The class A
complementary source follower pair will have less than 0.4% thd at full
power into 6 ohms, and
zero crossover distortion and Rout < 1 ohm, so without any global
NFB the output stage
is as good as many tube amps with 20dB of global NFB. The NFB is in the
source follower connection.
One could build a bjt driver amp as I have above with the 2 x
300w with low thd by having a loop of FB from
the MJE340/350 collectors back to the differential input amp NFB input.
The set up of the MJE340/350
gain pair would require that the NFB resistance network load the bjts
to reduce their open loop gain somewhat
so that a lower amount of NFB is actually used.
I am sure the
brighter mortals amoung you will tailor your diy efforts to suit
yourselves without smoke and
give your ears good music.
There are
1,001 ways to build an amplifier.
This shows the
inside of the 50 watt monos. The psu torodial power transformer
and choke are towards the left behind two 4,700uF caps, circuit board
in the centre,
more psu caps and 5Kg OPT with C cores at the rear right.